4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION
4.2 Novel transition concept with DMPA: First prototype in E-bandin E-band
4.2.2 Manufacturing and measurement of DMPA transitions at WR12
misalignment of the top mount of the PCB in x-axis and y-axis, respectively. The tolerance range is selected as 100 µm for both parameters. Simulated results are shown in Figure4.13.
The results show that the transition has robust performance for assembling tolerance.
Figure 4.11: Top view of the transition with gap coupled DMPA. c2010 IEEE [144].
Figure 4.12: Simulation results of the transition with gap coupled DMPA, w1=0.46 mm, w2=0.40 mm, w3=1.00 mm, w4=1.00 mm, l1=1.00 mm, l2=0.94 mm, g1=0.10 mm, g2=0.35 mm,s0=0.76 mm,s1=0.24 mm,s2=0.56 mm,s3=0.24 mm. c2010 IEEE [144].
4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION
Figure 4.13: Tolerance ofd1andd2 in gap-coupled DMPA transition.
Measurement step A: Two port measurement for back-to-back test boards Since the transitions have an asymmetric port structure – waveguide port and coupled microstrip line port – the most common way is the back-to-back (B2B) test structure. Both designs were first verified in back-to-back test structures. Figure 4.14 shows a photo of the manufactured top mount which is made of brass. The top mount was designed with WR12 compatible pin and screws. Figure 4.15 shows the PCB part of the back-to-back (B2B) test structure of both transitions.
Figure 4.14: Photo of top mount part of transition.
The measurements were performed by two-port vector network analyzer (VNA). Figures 4.16and4.17show the measurement results for both designs. There are some frequency shifts because of the manufacturing tolerance. The measurement of the second manufacturing of gap-coupled DMPA transitions B2B test board, as well as simulation results show good agreement (see Figure 4.18). From the B2B measurement, the functionality of the transitions has been 76
(a) B2B structure of the transition with single DMPA.
(b) B2B structure of the transition with gap-coupled DMPA.
Figure 4.15: Photo of test structure of transitions with single DMPA (a) and gap-coupled DMPA type (b). c2010 IEEE [144].
Figure 4.16: Measurement results of B2B structure of the transition with single patch DMPA.
c 2010 IEEE [144].
verified. The insertion loss as well as the return loss can be estimated. The ripples of return loss reduce the measurement accuracy. This is improved in measurement step B.
Measurement step B: One port measurements of test boards with spiral struc-ture
In step B, the transition is measured as one port device from waveguide port. The other port of transition – the coupled MSLs port – is terminated with spiral structure. In this step, gap-coupled transition with spiral structure at the coupled MSLs port has been manufactured and tested. The spiral structure, together with the absorber material on top, behaves as a good matching load. Figure4.19shows the measured return loss (waveguide port) of such test struc-ture with and without the absorber material. The return loss of transition without the absorber material has ripples in the measured bandwidth. The ripples are coming from the reflection 77
4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION
of coupled MSLs port. In the measurement of the transition with the absorber material, the ripples are gone. Therefore, the absorber material shows improvement for the matching load. A couple of differential absorber materials are tested. Among them, the best materials are QR13 and CRAM369. In general, the transition shows 10 dB return loss bandwidth from 74 GHz to 82 GHz. It matches the simulated results of the transitions. It is slightly larger than the simulation results because of loss in the substrate.
Measurement step C: LRdR for coupled MSL port measurement
Till now, the concept of the transition has been verified by insertion loss and return loss of the waveguide port. More difficult is the return loss of the coupled MSLs port under different modes. In this part, the author developed a LRdR (load, reflect, delayed reflect) method for verifying the return loss of the coupled MSLs port.
The transition can be treated as a three-port device, include asymmetric MSL port and
Figure 4.17: Measurement results of B2B structure of the transition with gap-coupled DMPA.
c 2010 IEEE [144].
Figure 4.18: Measurement results of GC DMPA transition on TLE95 material vs simulation.
78
Figure 4.19: Measurement results of transition with spiral load wt/wo absorber material.
waveguide port. It is possible to simplify the three-port device to a two-port device in the measurement. The solution is using three different terminators on waveguide port in three measurements. Hietala introduced a method for determining two-port S-parameters from only one-port measurement results in 1999 [146]. The same model can be extended to three-port device. The measurement setup of the transition can be described as a loaded two-port network.
Figure 4.20 shows a block diagram of a loaded two-port network and its equivalent S-parameter model. The transition is depicted as a device under test (DUT), which has two types of port. Port 1 is coupled MSL port, and port 2 is waveguide port. The matrix [S] represents the S-parameters of the transition. Standard connectors are connected to port 2. Γs,i denotes the reflection coefficient of the i-th standard connector. Γi is the reflection coefficient measured at port 1 with the i-th standard connector at port 2. It must be noted that there are two different modes of a coupled microstrip line (differential mode and common mode); therefore, Γi is different for each mode. The coupling between models are neglected here because of symmetric structures. Γi must be selected for the correct mode when extracting the S-parameter matrix of the transition.
Considering Figure 4.20, it is easy to deduce the relationship among Γi, Γs,i and [S] as follows
Γi=S11+S12·S21·Γs,i
1−S22·Γs,i (4.1)
Taking the reciprocal property of the transition into account (S12 =S21), there are three unknowns in Equation 4.1: S11, S12, andS22. Therefore, using the reflection coefficients Γs,i of the three standards, [S] can be calculated. Closed-form formulas of calculation are derived on the basis of matrix calculation.
Multiplying both sides of Equation4.1with (1−S22·Γs,i) results in
S11−S11·S22·Γs,i+S122 ·Γs,i+S22·Γs,i·Γi= Γi (4.2) 79
4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION
(a) Block diagram of measurement of LRdR. c 2011 IEEE [147].
(b) Signal flow diagram.
Figure 4.20: Block diagram of measurement and signal flow diagram.
To obtain a linear equation,W is introduced here:
W =−S11·S22+S122 (4.3)
Inserting Equation4.3into Equation4.2, a linear equation is reached for the three unknowns:
S11,W, andS22:
S11+W ·Γs,i+S22·Γs,i·Γi=·Γs,i·Γi (4.4) With three different standards – matched load, reflector, and delayed-reflector – we obtain solutions as follows:
1 Γs,M L Γs,M L·ΓM L
1 Γs,R Γs,R·ΓR
1 Γs,d−R Γs,d−R·Γd−R
S11
W S22
=
ΓM L
ΓR
Γd−R
(4.5)
S11,W andS22can be calculated as
S11
W S22
=
1 Γs,M L Γs,M L·ΓM L
1 Γs,R Γs,R·ΓR
1 Γs,d−R Γs,d−R·Γd−R
−1
ΓM L
ΓR
Γd−R
(4.6)
After obtaining the array, [S11, W and S22], S12 can be calculated according to Equation 4.3.
S12=±p
W +S11·S22 (4.7)
The sign ofS12 can be determined from prior information about the phase.
It is important to note that none of the above derivations depend on the port types of the DUT. The crucial point is that the Γs,i must be measured with the same port impedance of 80
port 2 of DUT. This method is suitable for measuring a transition with different types of port.
However, it is not limited to transitions; it can be used to extract the S-parameters of other hybrid 2-port device.
The test board, as well as measurement setup, is shown in Figure4.21. The design of gap-coupled transition is implemented as DUT. The test board has a waveguide port interface and a coupled microstrip lines interface. Because the coupled microstrip lines cannot be measured directly by G-S-G probe tips, we convert the coupled microstrip lines to two single-ended microstrip line ports. Those two single-ended MSL ports can be measured by G-S-G probe tips of a VNA.
Since open is not a good choice for waveguide port, in the project, the three standard terminators are matched load, short, and delayed short. This leads to the name LRdR (load, reflector and delayed reflector).
The measurement procedure consists of two steps. The first step is to measure the reflection coefficients Γs,i of the three standards. This is a one-port measurement of the waveguide port.
The second step is to measure Γi of coupled MSLs port with different standard connectors at the waveguide port. This is a two single-ended MSL ports measurement. Subsequently, the Γi
of two different modes can be calculated from these measurements.
Calculation steps:
The coupled microstrip line has two different modes: differential mode and common mode.
Therefore, the first step of the calculation is to convert the single-ended measurement results into a mixed-mode matrix according to the following equations in [110].
M = 1
√2
"
1 −1
1 1
#
(4.8)
SM M = M·SSE·M−1 (4.9)
whereSSEdenotes the two-port single-ended measurement result. SM M is the mixed-mode matrix, in which S11,M M stands for the reflection coefficient in differential mode, andS22,M M
stands for the one in common mode. Using this conversion, we obtain Γi of the two different modes.
The second step is to calculate the S-parameters of the transition using Γi and Γs,i. The model and formulas4.5,4.6,4.7are used here to calculate the S-parameters either in differential mode or in common mode.
The calculated results are plotted in Figure4.22. To provide a better comparison, the results include simulation results for a single transition and measurement results from a B2B structure of the transition.
Figure4.22(a)shows the return loss of the waveguide port from the simulation, the presented method, and the B2B structure. The bandwidth of the return loss from the B2B structure measurement is difficult to determine, since there were several ripples that distorted the results.
But from the presented method, the ripples are much more moderate. It is easy to determine that the 15 dB return loss bandwidth is 7.4 GHz, which matches the simulation results (6.6 GHz). The center frequency shift is caused by the fabrication tolerance.
Figure 4.22(b) shows the return loss of the coupled MSLs port, the differential-mode and common-mode return loss of the coupled MSLs port are plotted for comparison, including both 81
4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION
Figure 4.21: Photo of test board of GC DMPA transition in E-band for LRdR measurement setup. c2011 IEEE [147].
simulated and calculated results. It must be noted that these results cannot be obtained from any two-port measurement of B2B structures of the transition. The calculated results show that 15 dB return loss bandwidth of the differential mode is much wider than that of the common mode. This agrees with the simulation results. The narrow bandwidth of the common mode is caused by the parasitic patch within the transition. There are still some moderate ripples in the return loss because of the feeding lines.
The improvement that our method provides is even clearer in the calculated insertion loss (differential mode of coupled MSLs to waveguide) (see Figure 4.22(c)). The strong ripples in the B2B measurement are removed completely when using LRdR method. The higher loss in the measurement is coming from the fabrication tolerances and surface roughness.
Finally, in Figure4.22(d), the insertion loss of both differential mode and common mode is plotted together for comparison. Theoretically, the common-mode insertion loss will be very high, but because of fabrication and measurement limitation, it is about 6 dB lower than the differential-mode insertion loss. Taking the high reflection of the common-mode signal into consideration, the common-mode propagations are highly suppressed by the transition. This fulfills the design target.