Design of DMPA transition at 79 GHz

The transition has two parts: top mount (made of brass) and PCB part (made of RF substrate material). Here, the design for the top mount is shown first. The top mount is a modified waveguide line. In this work, waveguide size is selected as WR12 (a= 3.1 mm,b= 1.55 mm), 69

4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION

Figure 4.4: 3-D view of the transition structure.

which has an operating frequency range from 60 GHz to 90 GHz. The cross section of the transition is shown in Figure4.5.

Figure 4.5: Cross section of the transition (x-z plane). c2010 IEEE [144].

The width of the channel (W_c) is larger than the narrow wall (b) of the waveguide, while the height of the channel (Hc) needs some trade-off. IfHc is too high, there are risks of wave leakage from the top mount; ifHc is too low, the top mount will interfere the wave propagating along the coupled MSLs. In this design, Hc is selected as 1 mm, which is about eight times of the substrate thickness. It is an optimized value for the trade-off leakage suppression and high-order mode suppression [145]. It is worth to mention that the top mount part is linked to waveguide band. It can be reused in same waveguide band applications.

The key component of the transition is the DMPA part. The following part of the section gives two examples of the design. The top mount parts are the same and can be shared for both designs. The differences are the DMPA part designs. The first design is a single patch DMPA inside the transition, and the second is a gap-coupled (GC) DMPA inside the transition.

The RF substrate is selected as Taconic TLE-95 (εr=2.95, h=0.127 mm, tanδ=0.004). Here, designed details for both DMPA designs are given respectively.

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A. Single patch DMPA transition

The simplest way to build such a transition is to place a DMPA which has been verified in the previous chapter into a waveguide. Therefore, the first design is a single patch DMPA transition. The top view of the PCB is shown in Figure 4.6. The waveguide area is marked as rectangular shape with the size ofabyb, wherea= 3.1 mm andb = 1.55 mm. The patch antenna is placed in the center of the waveguide area. Since the patch size is smaller than the waveguide size (abyb), it is feasible to put a patch inside the waveguide. The radiating edges of the patch are in parallel with the broad walls of the waveguide. The feeding lines are from the narrow wall of the waveguide. In such configuration, the coupling between the patch and the waveguide are maximum maintained. A C-shaped vias array surrounds the patch antenna area.

The vias array is outside of the waveguide area. It is for suppressing surface wave propagation inside the substrate. The vias have a diameter of D=250µm and pitch of d=400 µm.

Figure 4.6: Top view of PCB part for single DMPA transition,a=3.1 mm,b= 1.55 mm,D=250 µm,d=400µm,W_c=7 mm. c2010 IEEE [144].

The DMPA is the key component of the transition. The main difference between the dif-ferentially fed antenna and the single-ended antenna is that the former one is fed through the non-radiating edge of the patch. Therefore, no inset at the radiation edge of the patch is necessary. Integrating a differentially fed antenna into a transition also removes the need for cutting into the long side walls of the waveguide. Hence, good coupling effects between the antenna patches and the waveguide walls are retained. Furthermore, the waveguide walls work as electric shield walls for the patch. Therefore, more fields are in free space compared with the classic microstrip patch antenna in open air. This reduces the Q factor of the patch. In the remainder of this section, the design procedures are shown in detail. Later on, the results are presented and discussed.

After building the first model, let us have a look at the difference to DMPA in open air. The waveguide can be simplified as shielded wall around the patch, together with vias array. Figure 4.7 shows the model of patch in open air and waveguide, anddis the distance from the patch edge to the waveguide wall. Figure 4.8gives simulation results for three different cases. The middle curve (blue) shows the patch in open-air situation, andd=inf ihere means the shielded 71

4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION

(a) E-field of microstrip patch antenna in open air.

(b) E-field of microstrip patch antenna with shielded wall.

Figure 4.7: Cross section (x-y plane) of E-field of microstrip patch antenna in open air (a) and with shielded wall (b). Top view of PCB part for single DMPA transition. c2010 IEEE [144].

wall has infinite distance to the patch. After adding the shielded wall, there are two kinds of effects. If the distance is relatively large, for instance, twice the thickness of the substrate (h), the resonant frequency shifts to the higher part (red curve). If the distance is relatively small, for instance, half of h, the resonant frequency will shift to the lower part (orange curve). In both cases, the relative bandwidth of the patch is bigger than the patch without the shielded wall.

Those effects can be explained by further study of the E-field distribution in the transition cross section. Figure4.7shows the cross-section E-field distribution of patch with and without the shielded wall. In an open-air situation (see 4.7(a)), the E-field is concentrated in the substrate. By adding shielded wall, the E-field is distracted by the wall. In other words, the E-field beneath the patch is getting less (see4.7(b)). This fact is responsible for high shifting the resonant frequency of the patch.

Figure 4.8: Simulated return loss of transition for differentd, where dis the distance from the patch edge to the waveguide wall.

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There is another effect while putting patch inside the waveguide, and that is the increasing capacitance between the edge of the patch and the shielded wall. It will become the dominant effect when the shielded walls are close to the patch edges, for instance, less than the thickness of the substrate (h). Therefore, bandwidth is increased by inserting the patch inside the waveguide.

Combining both effects, in most of the cases, because of manufacturing limitations, the spacing between the patch edges and the waveguide walls are larger thanh.

In the DMPA design procedure, the feeding position of the patch determines its impedance.

This also holds true for the transition design. A wider spacing in the DMSL corresponds to a higher impedance of the microstrip port.

Figure4.9shows the simulation results for different patch width-to-length ratio (W/L). The maximum bandwidth occurs when the W/L ratio is 1.1. It is different from DMPA in open-air design and keeps the DMPA part compact in transition design. Figure 4.9 also adds the bandwidth of DMPA in open air for comparison. The simulation results show that the relative bandwidth for 10 dB return loss of single patch inside the waveguide reaches 6.6% while in open air, it is about 4%.

Figure 4.9: Relative bandwidth of 10 dB return loss for the patch in waveguide.

Such a transition model was built in CST MWS and optimized for 78 GHz. Figure 4.10 shows the simulation results with optimized dimension. The coupled MSLs have a dimension ofWm= 260µm andSm= 240µm. The bandwidth for 15 dB return loss of the coupled MSLs port in differential-mode reaches 2.66 GHz, while 10 dB return loss is 4.5 dB. The common-mode signal is strongly suppressed for the whole frequency range because both the patch and the waveguide do not support the common-mode wave propagation. The common-mode return loss of the coupled MSLs port is less than 1 dB in whole E-band (60–90 GHz). The simulated insertion loss is about 0.3 dB. Theoretically, the common-mode return loss should be 0 since the structures are fully symmetric. But because of asymmetric structure in the realized examples, there are certain level common-mode signals observed in the measurement.

B. Gap-coupled patch DMPA transition for wider bandwidth

The second design is intended to increase the bandwidth of the transition. There is a common method in antenna design for bandwidth improvement – adding a parasitic patch. It 73

4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION

Figure 4.10: Simulation results of the transition with single patch DMPA, w = 1.10 mm, l= 1.02 mm,W_m= 0.26 mm,S_m= 0.24 mm. c2010 IEEE [144].

can be easily implemented in the transition design, especially with DMPA structure.

The top mount is identical as in the first design. The modification is only the DMPA part on PCB. The top view of the gap-coupled patch DMPA transition is shown in Figure4.11. In single patch DMPA transition design, the patch has a compact size (1.02 mm by 1.10 mm) – almost a square shape. It brings benefit for adding additional patch in the waveguide area.

The DMPA has two patches: the main patch and the gap-coupled (or parasitic) patch. The main patch is similar to the single patch DMPA. The parasitic patch is placed alongside the non-radiation edge of the main patch. The main patch is connected with coupled MSLs, and the parasitic patch is gap-coupled with the main patch.

With different dimensions of the two patches, two different resonant frequencies could be reached. The patch length of parasitic patchl2has a smaller size than the main patchl1. Since the length of patch determines the resonant frequency, the main patch corresponds to a lower resonant frequency while the parasitic patch corresponds to a high resonant frequency.

The differential signals are injected in the main patch through the coupled MSLs. Sub-sequently, through coupling effects, the electromagnetic power on the main patch excites the parasitic patch. Therefore, the bandwidth of the transition is almost doubled here. It is dif-ferent from the classic single-end MPA transition where the patch is center placed, and there is less space for a second patch within the waveguide area. After adding the second patch, the impedance matching cannot be reached without a matching network. The matching network is composed of two sections of transmission lines. The first one, from A-A’-plane to B-B’-plane, is shorter than a quarter of the wavelength. It converts the patch impedance into a value greater than 100 Ohm on the real axis of the Smith Chart. The second one, from B-B’-plane to C-C’-plane, has a high characteristic impedance. It works as a quarter-wavelength transformer and matches the 100 Ohm differential-mode impedance of coupled MSLs. The simulated S-parameters of the optimized GC DMPA transition are shown in Figure4.12. It shows 7 GHz bandwidth for 15 dB return loss which is more than double of single DMPA transition. The common mode return loss is slightly high at 78 GHz.

Assembling tolerances are studied by simulation for parameterd₁andd₂. d₁andd₂are the 74

misalignment of the top mount of the PCB in x-axis and y-axis, respectively. The tolerance range is selected as 100 µm for both parameters. Simulated results are shown in Figure4.13.

The results show that the transition has robust performance for assembling tolerance.

Figure 4.11: Top view of the transition with gap coupled DMPA. c2010 IEEE [144].

Figure 4.12: Simulation results of the transition with gap coupled DMPA, w₁=0.46 mm, w₂=0.40 mm, w₃=1.00 mm, w₄=1.00 mm, l₁=1.00 mm, l₂=0.94 mm, g₁=0.10 mm, g₂=0.35 mm,s₀=0.76 mm,s₁=0.24 mm,s₂=0.56 mm,s₃=0.24 mm. c2010 IEEE [144].