Differential feed antenna in millimeter wave Radar applications / submitted by Ziqiang Tong

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Submitted by Ziqiang Tong Submitted at

Institut f¨ur Nachricht-entechnik und Hochfre-quenzsysteme Supervisor and First Examiner Univ.-Prof. DI Dr. An-dreas Stelzer Second Examiner

Prof. Dr.-Ing. Wolfgang Menzel Co-Supervisor Name of assistant February 2020 JOHANNES KEPLER UNIVERSITY LINZ Altenbergerstraße 69 4040 Linz, ¨Osterreich www.jku.at DVR 0093696

Differential feed antenna

in millimeter wave Radar

applications

Doctoral Thesis

to obtain the academic degree of

Doktor der technischen Wissenschaften

in the Doctoral Program

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T o my parents,

my f ather M aoda T ong, and my mother Lihua J iang and my f amily,

my wif e J ing and my son M ichael.

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Acknowledgements

I would like to take this opportunity to thank all the people without whom this Doctor dissertation would not have been possible. I would like to thank my advisor, Prof. Dr. Andreas Stelzer, for all the opportunities you have provided for me to better understand the field of catalysis. I am also indebted to my co-supervisor Prof. Dr. Wolfgang Menzel for his guidance and understanding in the course of this program.

The research was made possible by the financial assistance from the Danube Inte-grated Circuit Engineering GmbH (DICE GmbH). I appreciate the cooperation and useful discussions of Dr. Eric Kolmhofer and Dr. Linus Maurer. I really appreciate all their help and brilliant ideas that helped channelize my research.

I am also grateful to Mr. Ralf Rudersdorfer and Mr. Johann Katzenmayer, for their assistance in the manufacturing process.

I thank all my committee members for their help, guidance, and knowledge you have provided me. I would not be where I am now without your questions, suggestions, and insights. I would like to thank my colleagues at the Institut fuer Nachrich-tentechnik und Hochfrequenzsysteme (NTHFS) and my research group members for their help and useful scientific discussions, especially, Dr. Reinhard Feger, Dr. Thomas Wagner, Dr. Martin Jahn, Dr. Alexander Fischer, Dr. Abouzar Hamidipour, Dr. Xin Wang, Dr. Christoph Wagner, etc.

Lastly, I would like to thank my family: Jing, my wife, and Michael, my son, for believing in me and supporting me.

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Abstract

This thesis presents a study of differential feed antenna in millimeter radar applica-tions. In recent years, millimeter-wave radar (30–300 GHz) has been exploited for a variety of applications, particularly in the automotive industry (76–81 GHz). A high-level integration of systems is desired to reduce the cost and achieve a com-pact size for mass production of radar systems. Most of the millimeter chips have differential topology and RF IOs, while the classic antennas are single-ended struc-tures. Therefore, there is a growing interest to develop a differential feed antenna for mmW radar systems. Differential antenna eliminates the need of a balun in the RF front-end system. This reduces the system size and the system loss. Meanwhile, the differential feed antennas also bring inherited benefits like lower cross polarization, etc.

The main focus of the thesis is to develop various differential feed antennas for mmW radar systems. Three groups of differential feed antennas have been studied. The first group are differential antennas on a microstrip structure. The microstrip structure is the most popular layer stack in mmW radar systems. In this part, the first differential feed microstrip patch antenna is designed based on a rectangular patch antenna. Then two different antenna arrays – H-plane array and E-plane array – are developed for increasing the gain of the antennas. The H-plane array provides wide bandwidth, while the E-plane array provides better radiation performance. In the second group, the differential feed antenna is implemented as radiation ele-ments in the transition devices which connect planar structures and air-filled waveg-uide structures. A couple of novel transitions are designed based on various differen-tial feed patch antennas. These transitions provide a smooth connection from planar structures (coupled microstrip lines) to vertical structures (rectangular waveguide structures). This facilitates the integration of a waveguide antenna – like horn antenna – in the radar front-end systems. Advances in 3-D printed technology development mean that this solution has more and more wide applications. The third group is the differential feed antenna integrated in package. Antenna in package has much higher integration levels compared with antenna built in printed circuit boards. The eWLB package is a promising package solution for mmW ap-plications. It brings new facilities for the antenna in package development. In this part, differential feed antenna concepts are further extended. Three types of dif-ferential feed antenna in package are designed in the eWLB package: folded dipole antenna, folded dipole antenna with cavity in PCB, and dual patch antenna. To improve the radiation performance, two dielectric lens – hemisphere lens and rod lens – which are mounted on top of the package are also developed and verified. As part of this study, all of the antennas/transitions are manufactured and tested. Measured results are presented and discussed, validating design and simulations. Meanwhile, the theories for antenna/transition measurement are also developed.

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The antennas are implemented in a couple of different systems. For instance, planar structures are suitable for middle- and long-range radar applications, while waveg-uide structures are good candidates for high-performance radar where long trans-mission lines are needed. The antenna-in-package solution is a promising candidate for ultrashort range (< 20 m) applications, etc.

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List of Figures

1.1 Radar basic form . . . 1

1.2 FMCW basic . . . 3

1.3 MIMO radar . . . 4

1.4 Long-range radar from Bosch . . . 4

1.5 Gauge radar from Siemens . . . 4

1.6 FMCW radar front-end block diagram . . . 5

1.7 Solver Selection . . . 6

1.8 Transient Solver . . . 6

2.1 Waveguide antenna in mmW applications . . . 8

2.2 Slot antenna on narrow wall of waveguide . . . 9

2.3 Spherical lens antenna . . . 9

2.4 Lens antenna fed by planar array . . . 10

2.5 Artificial lens at 76 GHz . . . 10

2.6 Lens antenna for 77 GHz: plano convex and planar lens . . . 11

2.7 Cylindrical parabolic reflector antennas . . . 12

2.8 Printed folded reflector antenna . . . 12

2.9 Open-ended transmission line antenna . . . 13

2.10 Differential fed grid antenna array on RO3003 . . . 14

2.11 Grid antenna on LTCC . . . 14

2.12 Series-fed patch antenna array . . . 15

2.13 Series-fed patch array in phase-shift receiver system . . . 16

2.14 Dual-fed phased array . . . 16

2.15 Dual linearly polarized microstrip patch antenna array . . . 17

2.16 Circularly polarized microstrip antenna array . . . 17

2.17 Patch antenna in Bosch automotive radars . . . 18

2.18 SIW antenna on flex substrate . . . 18

2.19 Slot-pair SIW antenna . . . 19

2.20 AoC at 77GHz . . . 20

2.21 A 79GHz LTCC radar front-end . . . 20

2.22 Wide bandwidth LTCC radar front-end . . . 21

2.23 AiP by QFN packaging . . . 22

2.24 AiP by eWLB packaging . . . 22

2.25 Examples of antenna in automotive radar . . . 23

3.1 Cross section of microstrip line structure . . . 24

3.2 Electric and magnetic field lines at low frequencies with static approximation . . 25

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LIST OF FIGURES

3.3 Microstrip patch antenna . . . 25

3.4 Magnetic wall of microstrip patch antenna . . . 27

3.5 Electric field and magnetic surface current distributions along the periphery for various modes of a rectangular microstrip antenna . . . 29

3.6 Differential-fed antenna in centimeter-wave applications . . . 31

3.7 Differential-fed antenna in mmW applications . . . 31

3.8 SMPA and DMPA configuration . . . 32

3.9 Normalized input resistance of DMPA – middle feed . . . 36

3.10 Resistance ratio between DMPA and SMPA – middle feed . . . 37

3.11 Antenna impedance of DMPA with middle feed . . . 37

3.12 DMPA with middle feed and edge feed . . . 38

3.13 Real(Z) of DMPA with edge feed . . . 40

3.14 Real(ZDM P A) of DMPA middle and edge feed . . . 40

3.15 Imag(ZDM P A) of DMPA middle and edge feed . . . 41

3.16 DMPA with middle feed MSL and edge feed MSL . . . 41

3.17 Coupled MSL structure . . . 42

3.18 Simulated differential mode characteristic impedance of coupled MSL at 79 GHz 43 3.19 Single patch DMPA simulated return loss (RL) in Smith Chart . . . 43

3.20 Simulated differential mode E-field distribution at 79 GHz . . . 44

3.21 Parameter study of single patch DMPA – Lp . . . 44

3.22 Parameter study of single patch DMPA – y1. . . 45

3.23 Relative bandwidth of DMPA and SMPA with different Wp/Lp ratio . . . 45

3.24 Simulated reflection coefficient for the optimized DMPA . . . 45

3.25 Single patch DMPA prototype for S-parameter measurement . . . 47

3.26 Simulated and measured reflection coefficient of DMPA . . . 47

3.27 Single patch DMPA prototype for far-field measurement . . . 48

3.28 Normalized radiation pattern of a single DMPA for E-plane (y-z) . . . 48

3.29 Normalized radiation pattern of a single DMPA for H-plane (x-z) . . . 49

3.30 DMPA H-plane / E-plane extension indication . . . 49

3.31 Four-element H-DMPA array structure . . . 50

3.32 Four-element H-DMPA array matching network design . . . 51

3.33 Four-element H-DMPA array final dimension . . . 51

3.34 Four-element H-DMPA array E-field distribution at 79 GHz . . . 51

3.35 Four-element H-DMPA array prototype for S-parameter measurement . . . 52

3.36 Four-element H-DMPA Array S-parameter measurement and simulation results . 52 3.37 Four-element H-DMPA array prototype for far-field measurement . . . 53

3.38 Photo of far-field measurement setup . . . 53

3.39 Normalized radiation pattern of four-element H-DMPA array for E-plane . . . . 54

3.40 Normalized radiation pattern of four-element H-DMPA array for H-plane . . . . 54

3.41 Normalized radiation pattern of four-element H-DMPA array for frequency squint in H-plane . . . 55

3.42 Three-element E-DMPA array Structure . . . 55

3.43 Three-element E-DMPA array matching network design . . . 56

3.44 Three-element E-DMPA array final dimension . . . 56

3.45 Three-element E-DMPA array E-field distribution at 77 GHz . . . 57

3.46 Three-element E-DMPA array S-parameter measurement and simulation results . 57

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LIST OF FIGURES

3.47 Three-element E-DMPA array prototype RO3003 . . . 58

3.48 Three-element E-DMPA arrray prototype for far-field measurement . . . 58

3.49 Normalized radiation pattern of a three-element E-DMPA array for E-plane . . . 59

3.50 Normalized radiation pattern of a three-element E-DMPA array for H-plane . . . 59

3.51 Normalized radiation pattern of the three-element E-DMPA array for frequency range 76 GHz to 79 GHz . . . 60

3.52 Cross section of wideband transition type 1 for DMPA measurement . . . 61

3.53 PCB part of the transition . . . 61

3.54 Electric field of tapered transition . . . 62

3.55 Simulation results of taper transition . . . 62

3.56 Cross section of wideband transition type 2 for DMPA measurement . . . 63

3.57 Photo of tapered transition prototype (shim type) . . . 64

3.58 Simulation and measurement results of the return loss of transition . . . 64

3.59 DMPA array application examples . . . 65

4.1 Classic single-ended transition prior art work . . . 67

4.2 Differential port transition prior art work . . . 68

4.3 E-field in rectangular waveguide – TE01 mode . . . 69

4.4 3-D view of the transition structure . . . 70

4.5 Cross section of the transition . . . 70

4.6 Top view of PCB part for single DMPA transition . . . 71

4.7 Cross section (x-y plane) of E-field of microstrip patch antenna in open air and with shielded wall . . . 72

4.8 Simulated return loss of transition for different d . . . 72

4.9 Relative bandwidth of 10 dB return loss for the patch in waveguide . . . 73

4.10 Simulation results of the transition with single patch DMPA . . . 74

4.11 Top view of the transition with gap-coupled DMPA . . . 75

4.12 Simulation results of the transition with gap-coupled DMPA . . . 75

4.13 Tolerance of d1and d2 in gap-coupled DMPA transition . . . 76

4.14 Photo of top mount part of transition . . . 76

4.15 Photo of test structure of transitions with single DMPA and gap-coupled DMPA type . . . 77

4.16 Measurement results of B2B structure of the transition with single patch DMPA 77 4.17 Measurement results of B2B structure of the transition with gap-coupled DMPA 78 4.18 Measurement results of GC DMPA transition on TLE95 material vs simulation . 78 4.19 Measurement results of transition with spiral load wt/wo absorber material . . . 79

4.20 Block diagram of measurement of LRdR . . . 80

4.21 Photo of test board of GC DMPA transition in E-band for LRdR measurement setup . . . 82

4.22 LRdR measurement results and simulation results of the transition . . . 83

4.23 Measurement results of GCP transition on RO3003 and TLE95 material . . . 84

4.24 Structure of vertical transition between rectangular waveguide and coupled MSLs – short-ended parasitic patch . . . 84

4.25 Top view of the PCB of the transition with common-mode suppression . . . 85

4.26 E-field distribution of transitions with common-mode suppression . . . 86

4.27 Simulated S-parameters of the transition with common-mode suppression . . . . 86

4.28 Photo of W-band transition with common-mode suppression . . . 87

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LIST OF FIGURES

4.29 Measurement and simulation results of B2B structure of the transition with

dif-ferent lengths . . . 87

4.30 Repeatability test of transition B2B structures . . . 88

4.31 Structure of transition with extended ground DMPA . . . 89

4.32 Cross-section of transitions comparison . . . 90

4.33 Enlarged details of PCB design in extended ground DMPA transition . . . 90

4.34 Simulated S-parameters of an extended-ground transition . . . 91

4.35 Measured S-parameters of the B2B structures of the transitions, classic and pro-posed . . . 91

4.36 Application of GCP transition in polarimetric mmW radar system . . . 92

5.1 The geometry of the aperture-coupled microstrip patch antenna with LTCC so-lution . . . 93

5.2 LTCC antenna designs at 77/79 GHz radar applications . . . 94

5.3 Antenna-in-package solution of superstrate structure . . . 95

5.4 Parasitic stacked patch antenna . . . 95

5.5 QFN packaging solution for AiP . . . 96

5.6 Radiation beam optimization of eWLB AiP by stack structure . . . 96

5.7 Comparison of standard WLP and fan-out WLP . . . 97

5.8 Schematic process flow for a fan-out wafer level package . . . 98

5.9 Cross section of AiP with MMICs in eWLB package . . . 98

5.10 Folded dipole and equivalent regular dipole . . . 99

5.11 Simulated S11 of folded dipole AiP in eWLB packaging . . . .100

5.12 Photo of manufactured folded dipole AiP . . . .100

5.13 Radiation pattern measurement of AiP configuration . . . .101

5.14 Measurement and simulated radiation pattern of folded dipole AiP . . . .102

5.15 Cross section of folded dipole plus cavity in PCB . . . .103

5.16 Top view of folded dipole AiP with cavity simulation model and simulated an-tenna impedance . . . .103

5.17 Simulation model and simulated S11 of the AiP - FD with and without cavity . .104

5.18 Photo of the fabricated package of folded dipole with cavity on PCB and test board . . . .104

5.19 Measurement and simulated radiation pattern of folded dipole AiP with cavity at 76.5 GHz . . . .105

5.20 EIRP of AiP – folded dipole with cavity in PCB . . . .105

5.21 Cross section comparison of AiP with eWLB package and superstrate structure .106 5.22 Simulation model of AiP – dual patch . . . .107

5.23 E-field distribution of the AiP – DP for differential signal . . . .107

5.24 Antenna impedance of AiP – DP . . . .108

5.25 Simulated return loss of the AiP DP . . . .108

5.26 Bottom view of the AiP DP package . . . .109

5.27 Power measurement of the AiP DP . . . .110

5.28 EIRP of AiP DP . . . .110

5.29 Measured and simulated radiation patterns of the AiP DP at 76.5 GHz . . . . .110

5.30 Radiation pattern of the primary antenna with different mold size . . . .111

5.31 Displacement currents of the molds with different dimensions . . . .112

5.32 Cross section of hemisphere lens on eWLB AiP . . . .113

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LIST OF FIGURES 5.33 Simulated radiation pattern of hemisphere lens on eWLB AiP – FD with cavity .113

5.34 Photo of eWLB AiP test board with hemisphere lens . . . .114

5.35 Measured and simulated radiation pattern of hemisphere lens on eWLB AiP – FD with cavity . . . .114

5.36 EIRP of hemisphere lens on AiP . . . .114

5.37 Cross section of the eWLB package and the rod antenna . . . .115

5.38 Simulated gain of the AiP for different heights of the rod lens . . . .116

5.39 Simulation of S11 with the enhanced model . . . .116

5.40 Photo of AiP-DP test PCB with rod lens . . . .117

5.41 Photographs of the measurement setup in the absorber chamber and the AiP with lens mounted . . . .117

5.42 Measured EIRP of AiP with and without lens . . . .118

5.43 Simulated and measured gain of AiP with and without lens . . . .118

5.44 Measured and simulated radiation pattern of AiP rod lens at 78.3 GHz . . . .119

5.45 Measured and simulated radiation pattern of AiP rod lens at 72 GHz . . . .119

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Chapter 1 Introduction

This chapter first introduces the basics of radar and its application in millimeter-wave opera-tions. Later, the motivation of the thesis and the chosen simulation tools are shown. In the last section, the whole structure of the thesis is presented.

1.1 Radar

Radar is acronym for radio detection and ranging. It is used to locate distant objects by sending out radio waves and analyzing the echoes that return. Radar can detect the range, the speed, the angle, and even the shape of the object.

The first radar was invented by Christian Huelsmeyer in 1904 [1]. In this patent, he intro-duced for the first time an apparatus for detecting a distance object using electromagnetic wave (EM wave). Since then, many radar systems have been developed and implemented in many military and civil applications.

Figure 1.1: Radar basic form.

Figure 1.1 shows the radar basic form. A radar system can be simplified into two parts: (1) baseband block and (2) radio frequency (RF) front-end block (including antenna). The baseband block defines the waveform of transmitting radio signals and analyzes the received 1

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1. INTRODUCTION

signals. The RF front-end generates, transmits, and receives the radio signal. The antenna is the interface of EM wave propagation between the circuit and free space. The EM wave is transmitted by the transmitting antenna (Tx antenna), and the reflected wave is received by the receiving antenna (Rx antenna). The target information is calculated by comparing the received signal (Srx(t)) and the transmitted signal (Stx(t)) in time, frequency, and phase domain, etc.

Different radar systems implement different topologies for target detection. For instance, the pulse radar detects the range of the target by measuring the time delay of the echo signal. The continuous-wave radar (CW radar) detects the velocity of the target from the doppler shift of the received signal. The frequency-modulated continuous-wave (FMCW) radar has the advantage of both range and speed detection. Therefore, FMCW radars are widely used in many applications, such as the automotive radar.

1.1.1 FMCW radar

FMCW stands for frequency-modulated continuous-wave. From the Institute of Electrical and Electronics Engineers (IEEE) Standard Radar Definitions (686-2008), frequency-modulated continuous-wave radar is a radar transmitting a continuous carrier modulated by a periodic function such as a sinusoid or sawtooth wave to provide range data.

This section shows the basic principle of FMCW radar [2]. Figure1.2shows a brief explana-tion about FMCW radar. The transmitting signal (Stx(t)) is a frequency-modulated triangular wave (solid black line). The reflected signal (Srx(t)) (dashed red line) from the target is par-tially mixed with the transmitted wave. In this example, the target is indicated as approaching the radar.

The time delay (delta T) between Stx(t) and Srx(t), as well as doppler frequency, maps to the frequency of the beat signal (LO signal). They have the following relationship:

fBU= fR− fV (1.1a)

fBD= fR+ fV (1.1b)

where fR and fV are the range frequency and velocity frequency, respectively.

The positive sign in the formula represents the frequency of the beat signal (downbeat fBD) obtained where the transmitter frequency falls. The negative sign represents the frequency (up-beat fBU) of the beat signal obtained where the transmitter frequency rises. More descriptions of the FMCW radar are shown in the [2]. If the target is moving away from the radar, the sign selection in Eq. 1.1will be opposite.

1.1.2 MIMO radar

From the antenna design point of view, the angle resolution is defined by the aperture size of the antenna. Since multiple-input and multiple-output (MIMO) configuration may exceed the antenna aperture over the physical aperture size by multiplexing Tx and Rx channels, it greatly improves the angular resolution of the radar system, or in other words, it reduces the antenna size for a given angular resolution in a system. In addition, in most radar systems, digital beamforming (DBF) technologies have been widely implemented [3,4,5,6]. It enables fast scanning compared with physical beam scanning. With MIMO and DBF, the radar size 2

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Figure 1.2: FMCW basic. In this example, the target is indicated as approaching from the eco radar.

becomes more compact, and the total cost is also largely reduced. Both enable the radar in wider applications. Figure 1.3 shows the classic configuration of the MIMO antenna array in which two Tx antennas and three Rx antennas are included. The antenna aperture can be derived from the convolution of the two individual aperture distributions [7]. Three Rx antennas are placed with uniform spacing d2, while two Tx antennas are placed with spacing d1 in between them. The typical value of d1 is λ/2, where λ is the free space wavelength of the operating frequency to compromise angular resolution and grating lobes in antenna arrays. The distance between Tx (d1) is usually a multiple of the distance between Rx (d2). In this configuration, the virtual Rx antennas are realized between Tx1 and Tx2. The number of Rx antennas is effectively doubled so that the radar performance in terms of angle detection is improved.

In MIMO systems, coherent signals are transmitted through Tx antenna 1 and Tx antenna 2. The Tx signal between the channels are orthogonal either in time domain or in frequency domain, etc. The reflected signals from the target reach different Rx channels with different time and phase. DBF technologies enable high angular resolution based on those time and phase information. The spacing between Tx and Rx channels varies widely. For instance, compact Tx channels’ spacing may be combined with sparse Rx channels’ spacing, etc. More detailed discussions are referred to [8, 9].

1.1.3 Radar in millimeter-wave applications

Radar, especially millimeter-wave radar (mmW radar), has a wide range of civil applications, including automotive applications [10] [11]. It has been used in the automotive industry since the 1970s [12], [13]. After more than three decades of development, it has become a key for accident-free driving and autonomous driving in the future. The frequency regulation of automotive applications includes 24 GHz and 77/79 GHz for long range. More and more development are focusing on 77/79 GHz radar systems. Figure1.4shows a typical radar module in automotive applications [14]. It supports driver-assistance functions, such as autonomous cruise control (ACC), etc.

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1. INTRODUCTION

Figure 1.3: MIMO radar.

Figure 1.4: Long-range radar from Bosch: LRR3 c 2011 IEEE [14].

Figure 1.5: Gauge radar from Siemens. c 2003 IEEE [15].

Another example is the tank level gauge with a 24 GHz FMCW radar system [15]. Figure

1.5 shows the basic configuration of the level gauge radar. It measures the liquid level inside the tank. Higher operating frequency improves the range accuracy of the radar.

1.2 Motivation

This section explains the motivation for the development of differential feed antenna in a millimeter-wave (mmW) radar system.

First and foremost, the motivation is the system integration within silicon monolithic mi-crowave integrated circuits (MMICs). Silicon technologies strongly drive mmW radar develop-ments [16]. It enables high integration in the radar front-end and low cost of radar systems. A simplified radar front-end MMICs block diagram can be demonstrated as Figure 1.6. It in-cludes a voltage-controlled oscillator (VCO) for RF signal generation, power amplifier (PA) for Tx channel, and mixer for Rx channel.

In all of those MMICs blocks, differential topologies are widely implemented. For instance, reference [17] demonstrates VCO design, reference [18] demonstrates differential PA design, reference [19] shows mixer circuits, many works are reported for the whole transceiver (TRx) [20, 21, 22, 23], etc. All those differential topologies MMICs have some common advantages such as [24]

• wide swing range, 4

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Figure 1.6: FMCW radar front-end block diagram.

• rejection of common mode (CM) noise, • DC offset reduction, and

• zero IF (direct conversion).

Besides those, differential topologies also benefit layout design, for example, neglect the ground routing as in single-ended topologies. The radio frequency input/outputs (RF IOs) are preferred to be differential, while the traditional antennas for mmW radar are single-ended. It is natural to implement differential feed antenna in mmW radar systems. This removes the balun for compact systems and reduces the transmission loss between MMICs and antennas.

The second reason is that the differential feed antenna has superior radiation performance, for instance, lower cross-polarization, etc.

Last but not least, the differential feed antenna introduces an additional option for mul-tichannel integration. Thus, it supports highly integrated multichannel radar. The balun structure from the classic system is eliminated.

1.3 Simulation tool

The simulation tool used in this work is Computer Simulation Technology Microwave Studio (CST MWS). Its transient solvers are suitable for analyzing the wide frequency behavior of the devices with less port numbers (see Figure1.7).

CST MWS is a general-purpose electromagnetic simulator based on the finite integration technique (FIT) first proposed by Weiland in 1976/1977 [25]. This numerical method provides a universal spatial discretization scheme applicable to various electromagnetic problems ranging from static field calculations to high-frequency applications in time or frequency domain [26]. FIT discretizes the integral form of Maxwell’s equations rather than the differential one.

Transient Solver :

The CST MWS transient solver allows the simulation of a structure’s behavior in a wide frequency range in just a single computation run. Consequently, this is an efficient solver for most driven problems, especially for devices with open boundaries or large dimensions.

The transient solver is based on the solution of the discretized set of Maxwell’s grid equations. 5

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1. INTRODUCTION

Figure 1.7: Solver Selection (Courtesy of CST AG, Darmstadt, Germany).

To better understand the explanations, let us look at how the transient solver calculates S-parameters. The transient solver operates with time pulses, which can be easily transformed into the frequency domain via a fast Fourier transformation (FFT). The S-parameters can then be derived from the resulting frequency domain spectra:

Figure 1.8: Transient Solver. [26]

For instance, a division of the reflected signal by the input signal in the frequency domain yields the reflection factor S11. Within just one simulation run in time domain, the full broad-band information for the frequency broad-band of interest can be extracted without the risk of missing any sharp resonance peaks. It is very efficient for wide bandwidth design. CST MWS is a pop-ular simulation tool in mmW antenna design and is used in this project for electromagnetic (EM) simulations. More details are referred to [26].

1.4 Thesis structure

This dissertation has six chapters in total. Chapter 1 is the introduction (this chapter). Chapter 2 gives a brief view of the state of art of mmW antenna development.

Chapters 3 to 5 explain the different feed antennas in planar form, waveguide integrated form and antenna in package form, respectively. Chapter 3 discusses the differential feed an-tenna in microstrip structures. It first analyzes a single patch feed by differential signal. Then the antenna design extends to E-plane arrays and H-plane arrays. Chapter 4 discusses the dif-ferential feed antenna integrated with air-fill waveguide structure. A couple of wide bandwidth 6

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designs are discussed. Chapter 5 discusses the differential feed antenna in package. A novel fan-out package technology is implemented in the designs.

Lastly, Chapter 6, presents the conclusion and future development of differential feed an-tenna.

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Chapter 2 State of Art for mmW Radar

Antenna

This chapter provides an overview of the state of art for antenna in mmW radar applications. Automotive radar is the most popular application of mmW radar, and the examples in the chapter are mainly for these applications [7, 27]. At the end of the chapter, some antennas from realized systems are shown as examples.

2.1 Waveguide antenna

Waveguide antenna is a traditional antenna that has also been developed in many radar appli-cations. It offers advantages because of its mechanical stability and high gain property. Horn antenna is the most popular waveguide antenna, but it is bulky in size. In mmW radar systems, because of its low-profile property, slot waveguide antennas attract more interest. Prof. Ando presented a couple of high-gain waveguide antenna designs in [28]. Figure2.1presents a couple of different solutions, including cophase feed, alternating phase feed, radial line slot antenna, post-wall waveguide, etc. They are all good candidates for high-gain antenna designs. Prof. K. Sakakibara proposed slot antenna on the narrow wall of a waveguide with an alternative feed mechanism for the grating lobe suppression in [29]. It is a new way for MIMO antenna array configuration since each array of slot has narrow width (see Figure 2.2). In addition, unlike

Figure 2.1: Waveguide antenna in mmW applications. Copyright c 2010 IEICE [28] 8

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classic metal waveguide antenna, the manufacturing of antenna was done by metal injection molding for cost reduction. It provides another possibility for reducing the cost of waveguide antenna.

Waveguide antennas offer great advantages because of their low-loss transmission line, which brings higher efficiency for the antenna. But because of cost and manufacturing difficulties, these antennas have limited applications in the mass production of radar systems.

Figure 2.2: Slot antenna on narrow wall of waveguide. Copyright c 2000 IEICE [29]

2.2 Lens antenna

Lens antenna is another type of antenna with a long history. Spherical lens is a classic design of lens antenna. Spherical lens antenna is based on the refraction of electromagnetic waves at the

(a) Schematic of the spherical lens antenna system [30].

(b) Wide scanning array with 33-beam [31].

Figure 2.3: Spherical lens antenna: (a) schematic of single antenna and (b) photo of scanning array. c 2002 IEEE [31]

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2. STATE OF ART FOR MMW RADAR ANTENNA

(a) Photograph of radar system circuit boards. (b) Near-field behavior of the lens used in the system at 77 GHz.

Figure 2.4: Lens antenna fed by planar array. c 2014 IEEE [33]

lens’ surfaces (in the case of isotropic homogenous lenses) or within the lens’ dielectric material in the case of nonuniform refractive index lenses [32]. Schoenlinner presented a dielectric spher-ical lens antenna for 77 GHz automotive radar applications in [30,31], first for a single antenna, and later, he extended it to a wide scanning array. The feeding antenna is a finline tapered-slot antenna. Figure2.3shows a schematic layout and a photo of the manufactured samples. The scanning capability was realized by multiple feeding antenna for different directions. It requires many Tx/Rx channels to cover a large field of view (FoV).

Dielectric lens may combine with multiple Tx/Rx antennas [35]. Such configuration gen-erates multiple beams within detection range. The target angle information can be calculated by comparing the phase and the magnitude information between the beams. It is a promising solution for long-range radar applications, which requires approximately 20 degrees field of view (FoV). In [33], Lutz extended the concept by adding elevation angle scanning. The Tx and Rx

(a) Sketch of the internal view of double-focused lens.

(b) Photo of a foam-made double-focused lens with horn primary antenna source at 76 GHz.

Figure 2.5: Artificial lens at 76 GHz. c 2003 IEEE [34] 10

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antennas are placed in different vertical positions (see Figure2.4(a)).

Gallee presented another type of lens antenna—an artificial lens—which is composed of a group of parallel-plate waveguides in [34]. In this example, the artificial lenses consist of stacked parallel-plate waveguides of various lengths. The shape and length of the plate and the distance between the plates are the degrees of freedom for lens design. Unlike the dielectric lens, the equivalent refractive index of the artificial lens is smaller than 1. Figure2.5shows a foam-made double-focused lens with horn primary antenna source at 76 GHz.

Some other lens antennas have been presented by Prof. Chen’s group in 2015 (see Figure

2.6). In [36], a plano convex lens antenna was designed for 77 GHz, while in [37], the plano convex lens was replaced by a planar lens. These works presented different lens antennas as well as a combination with substrate integrated waveguide (SIW) antenna.

In general, lens antennas are bulkier compared with planar antennas. They also have a limitation of mounting position in automobiles.

(a) Photo of plano convex lens for 77 GHz [36]. (b) Photo of planar lens for 77 GHz [37].

Figure 2.6: Lens antenna for 77 GHz by Prof. Chen’s group: (a) plano convex lens c 2015 IEEE [36] and (b) planar lens c 2015 IEEE [37].

2.3 Reflector antenna

A reflector antenna uses either planar shape or other forms of metal to reflect electromagnetic waves. It has been used since the discovery of electromagnetic wave propagation in 1888 by Hertz.

The most popular type reflector antenna is the parabolic reflector antenna. The research group at Karlsruhe Institute of Technology (KIT) has developed a couple of different reflector antennas for automotive radar sensors. Park presented an offset solution for cylindrical reflector antenna fed by waveguide Luneburg lens in 2003 [38], while Beer presented a more compact solution with high-integrated Yagi-Uda antenna as source antenna [39]. Figure 2.7 shows the principle and photo of both solutions. Parabolic antenna has a sophisticated theory for design but requires high accuracy for manufacturing in mmW applications. In addition, it requires high maintenance for assembling in cars.

In 1999, Prof. W. Menzel introduced another type of reflector antenna–printed folded reflector antenna for 77 GHz automotive radar [40]. This type of antenna uses two printed 11

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(a) Side view of the cylindrical reflector antenna fed by waveguide Luneburg lens [38].

(b) Cylindrical parabolic reflector with Yagi-Uda an-tenna [39].

Figure 2.7: Cylindrical parabolic reflector antennas from the research group at KIT. c 2003, 2003 IEEE [38,39]

substrates to build a polarized grid on top and a twist reflector on the bottom. With adjusting the twisting and focusing requirement, the overall plane waves are focused and passed to the top grid plane. The antenna has good radiation performance. It supports very narrow half-power beam width (HPBW = 2.7 degrees) and low side lobe level (SLL = 24 dB). Folded reflector antennas have been successfully implemented in series productions. Figure2.8 shows the layout of the reflector. The scanning capability can be realized by mechanical scanning method—tilting the reflector plane. In 2001, MA-COM demonstrated a convex plane design for folded reflector antenna [41]. It increases the detection angle of the radar but introduces extra manufacturing challenges.

Figure 2.8: Printed folded reflector antenna by Prof. Menzel. c 1999 IEEE [40].

Like lens antennas, reflector antennas are good candidates for high gain antenna solution, but they are bulky and have high demand of manufacturing.

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2.4 Planar antenna

Planar antennas are the most popularly implemented antenna when realizing mmW radar systems because of its low profile and low cost. There are many types of planar antennas – such as wire antenna, grid antenna, patch antenna, etc. – which have been developed for mmW radar applications in the last two decades. In the following part of this subsection, the author gives a number of examples for different types of planar antennas.

2.4.1 Open-ended transmission line antenna

Open-ended transmission line can be implemented as a radiating element for antenna designs. It is easy to realize different polarizations by tilting the line angle. The application of this antenna in automotive radar was first reported by Toyota in 2000, to the best knowledge of the author. Iizuka from Toyota reported his work on a 45-degree polarized wire antenna (open-ended transmission line) in [42] and [43].

Figure 2.9 shows the configuration of the proposed antenna by Iizuka. The antenna is a series fed by microstrip line. The radiating elements – open-ended half wavelength (λr/2) microstrip lines – are placed alternatively on either side of the feeding line. The separation of the wire element is λr/2. The connection is direct coupling. Unlike the classic comb wire antenna, the proposed wire antenna has 45-degree polarization for reducing the interference between the incoming cars.

(a) 45-degree polarized wire antenna configuration. (b) Photo of a 45-degree polarized wire antenna ar-ray.

Figure 2.9: Wire antenna (open-ended transmission line) proposed by Toyota. Copyright c 2002 Toyota CRDL [43]

2.4.2 Grid antenna

Grid antenna is another type of planar antenna. It has a periodic rectangular loop structure. Each rectangular loop integrates the connecting line (long side of the loop) and radiating line (short side of the loop); thus, the total antenna is formed in a very compact way. Grid antennas date back to 1964 and 1981. Recently, grid antennas were extensively investigated by Prof. Zhang [44] in mmW applications. The research group from Ulm University presented a couple 13

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of designs in grid antenna at 79 GHz for automotive radar applications. Frei presented a grid antenna based on soft substrate—RO3003 in [45]. The antenna has a novel feeding structure with differential input signals. Figure 2.10shows the antenna configuration and photo of the manufactured sample.

(a) Grid antenna array configuration. (b) Photo of grid antenna on RO3003 with waveg-uide feeding.

Figure 2.10: Differential fed grid antenna array on RO3003 (h = 256 µm). c 2011 IEEE [45]. Bauer demonstrated another grid antenna design based on a multilayer structure – low-temperature co-fired ceramic (LTCC) substrate. A system solution of radar front-end is also shown in [46]. The antenna was designed as a microstrip structure by back fed of coaxial structure. Two antennas make a subarray pair to support a stable radiation pattern in broadside direction over a wide frequency bandwidth. The feeding network implements a laminated waveguide (LWG) structure, which gives low transmission loss compared with a microstrip line. Figure2.11shows the photo of the RF front-end as well as the antenna configuration.

(a) Photo of RF front-end built on LTCC material. (b) Top view (top) and cross section (bottom) of grid antenna array on LTCC.

Figure 2.11: 79 GHz radar front-end with grid antenna on LTCC: (a) photo of front-end and (b) antenna structure. c 2013 IEEE [46].

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2.4.3 Patch antenna

Patch antennas are the first structure to be introduced for the microstrip antenna and are still the most popular structure in radar applications. A lot of research has been done on patch antenna for automotive radar applications in the last two decades.

Series-fed patch array is the most common configuration of microstrip antenna array. Schoebel summarized a couple of designs for series-fed patch array in [47]. Those antennas include uni-form array, amplitude-tapered array, amplitude-tapered inclined array, phase and amplitude optimized and high-gain array, etc. (see Figure 2.12). In some other designs, the Wilkinson dividers in high-gain antenna were replaced by T-junction dividers [12,48].

Figure 2.12: Series-fed patch antenna array examples: (a) patch column with uniform series feed, (b) amplitude-tapered patch column, (c) amplitude-tapered inclined patch column, (d) phase and amplitude optimized column and (e) high-gain antenna array – 8-column/12-patch array using Wilkinson dividers with mounted resistors. c 2012 Schoebel J, Ituero Herrero P. Published in [47] under CC BY 3.0 license.

Since MIMO configuration is becoming more and more popular in radar applications, the research group at Toyota developed a 16-patch series-fed array and applied it in a phase-shift receive system [49,50]. It implements tapered patch width for side lobe level optimization. The separation between the arrays are 0.6λ0, where λ0 is the wavelength of operating frequency in free space. The photo of the manufactured antenna and layer stack is shown in Figure 2.13.

To realize elevation scanning capability, a novel concept for adjusting antenna beam has been proposed by Topak [51]. In this work, the classic series-fed patch array was fed from both ends of the array (see Figure 2.14). The radiation beam can be controlled by tuning the amplitude and phase of the two feeding signals. It introduces a new solution for elevation information detection for automotive radar applications.

Many other types of patch antennas have been tried for automotive radar application re-cently. Shin developed an inclined antenna array by combining gap-coupled patch and direct-coupled patch [52] for wide bandwidth. Dewantari proposed a novel design for side lobe sup-pressing by a complementary split ring resonator (CSRR) structure [53]. Hamberger from the Technical University of Munich (TUM) presented new designs for different polarizations of patch antenna in the 77 GHz band [54,55]. Reference [54] shows a dual-polarized patch antenna array which is a potential candidate for polarimetric radar. The radiating element in this design is a 15

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(a) Phase array Rx antenna - 16 series-fed arrays. (b) Layer stack-up of the Rx antenna board.

Figure 2.13: Series-fed patch array in phase-shift receiver system. c 2013 IEEE [49].

(a) Block diagram of the dual-fed phased array pro-totype.

(b) Photograph of RF board employing a linear ar-ray antenna and MMIC phase shifters on the mea-surement platform.

Figure 2.14: Dual-fed phased array block diagram (a) and photo of RF board (b). 2013c IEEE [51].

square patch which is fed by two MSLs from the adjacent edges of the patch (see Figure2.15). Each feeding line is coming from a separate channel of MMICs. In [55], Hamberger extended the design to circular polarization antenna. The two feeding lines were replaced by a single feeding line connected with a power divider at a 90-degree phase offset (see Figure2.16). The challenge to these proposals is using very narrow MSL (width<0.1 mm) within the design. For this, a laser etching system was implemented in manufacturing. The mass production methods are still under investigation.

Xu from Southeast University (SEU) introduced a new Tx antenna design for combining long-range and middle-range applications in one single antenna [56, 57]. The antenna beam patterns were optimized as a shoulder shape – high gain in the broadside direction and middle gain for the off-broadside direction. The power distribution parts were based on substrate inte-16

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(a) A single element of the array column including the feed network.

(b) Antenna array model designed on a RO3003 sub-strate.

Figure 2.15: Dual linearly polarized microstrip patch antenna array: (a) single element model and (b) antenna array model. c 2016 IEEE [54]

(a) Simulation model in CST microwave studio. The splitter has an additional quarter wave section (90-degree phase shifting) in up branch of the feeding lines.

(b) Photo of the array with measurement setup.

Figure 2.16: Circularly polarized antenna array: (a) simulation model and (b) photo of the array. c 2017 IEEE [55]

grated waveguide (SIW) structures, which have less radiation loss than microstrip structures. The proposed antennas were verified by different RF substrate materials such as Taconic TLY-5 [57] and Rogers RO3003 [56], respectively. It shows a novel configuration of the antenna beam pattern while the system implementation is under development.

There are many patch antenna/series-fed arrays which have been implemented for automo-tive radar mass production (see Figure2.17). For instance, in Bosch long-range radar (LRR3), four rectangular patches are implemented as source for the lens antenna. There are two side patches for bandwidth enhancement. In middle-range radar (MRR), two Tx and four Rx an-tennas are all series-fed patch array. One of the Tx antenna uses a short pitch between the patch (< λr/2) for shifting the maximum radiation beam from the broadside.

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(a) Bosch long-range radar (LRR3). (b) Bosch middle-range radar (MRR).

Figure 2.17: Patch antenna in Bosch automotive radars.

2.4.4 Substrate integrated waveguide (SIW) antenna

Another type of planar antenna is substrate integrated waveguide (SIW) antenna. The SIW structure was first promoted by Deslandes in 2001 [58]. It integrates waveguide structure on a soft substrate material. The broad walls of waveguide are formed by top and bottom metals, while the narrow walls are formed by metallized vias array or grooves. Because of its low-profile and low-cost (compared with air-filled waveguide) property, many SIW antennas have been studied for automotive radar applications.

The classic SIW antenna uses slot as a radiating element. The slots are in parallel with the longitudinal axis of the waveguide, in other words, the propagation direction of the waves. The radiating slots are placed on the broad wall of the SIW—top side. Cheng proposed such a design of SIW antenna based on a flexible substrate – Kapton HN polyimide foil [59]. It shows a potential solution for mounting radar module on a convex surface (see Figure 2.18).

(a) Top view (up) and side view (down) of the single-column SIW antenna.

(b) Photograph of the folded SIW-based 4 by 4 slot array antenna.

Figure 2.18: SIW antenna on flex substrate: (a) top view and side view of the antenna config-uration and (b) photo of the manufactured folded SIW antenna. c 2009 IEEE [59].

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(a) Top view (up) and cross section (down) of slot-pair SIW antenna.

(b) The photo of the fabricated 4-column high-gain antenna array.

Figure 2.19: Slot-pair SIW antenna at 77 GHz: (a) sketch of single antenna array and (b) photo of the four-column array. c 2014 IEEE [60].

Prof. Wang presented another design of SIW antenna based on a RO5880 substrate [60]. The radiating elements are slot-pair, which are perpendicular to the longitudinal axis. The whole antenna is composed of four columns of arrays, and each array has 22 slot-pair elements (see Figure2.19). Massen designed a 3×15 subarray of SIW antenna at 79 GHz and implemented it into a MIMO radar design [61]. In the design, the slot widths are optimized for side lobe level. The SIW antenna is a good candidate for mmW radar antenna and has been implemented in realized radar systems.

2.5 High-integration antenna

There is another trend of antenna development for high-integration antenna. A lot of research work has been done in mmW radar applications, such as antenna on chip (AoC), antenna in package (AiP), etc. This section gives a short introduction for such type of antennas.

2.5.1 Antenna on chip

To the best of the author’s knowledge, the first publication of antenna on chip (AoC) at 77 GHz applications has been reported by Babakhani in 2006 [63]. It has been addressed that the AoC has very low efficiency compared with classic antennas on PCB since the substrate layer is very thin. In [63], Babakhani proposed a solution to radiate from the bottom of the silicon and combine it with the lens.

Hasch reported a novel solution of AoC for 77 GHz [62]. In his works, a parasitic resonator element is added on top of the patch antenna on silicon to increase antenna efficiency (see Figure

2.20). The parasitic resonator element is formed by a quartz glass whose length is equal to half the wavelength size. Hasch further investigated the system performance of AoC in [64]. In [64], AoC was integrated in LRR module by replacing the original source antenna—patch antenna. The system performances were measured and compared with LRR module. In general, AoC has high integration level but low radiation efficiency and high cost.

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(a) Photo of AoC with parasitic resonator element on complete transceiver.

(b) Photo of AoC in LRR3 module housing.

Figure 2.20: AoC proposal by Bosch: (a) photo of AoC and (b) AoC in LRR3 module. c 2010 IEEE [62].

2.5.2 Antenna in package

Antenna in package (AiP) is another solution for high-integration system. It balances the cost and efficiency between antenna on chip and antenna on PCBs. It is getting more and more attention in the mmW radar development.

There are also some recent works on multilayer structure antenna.

Vasanelli presented a multilayer structure, aperture-coupled antenna design based on Rogers material [65]. Mosalanejad showed another proposal for multilayer structure [66].

Besides the soft substrate, because of its low-loss and high dielectric constant property, low-temperature co-fired ceramic (LTCC) is getting more and more attention in automotive radar research studies. X. Wang presented a sophisticated solution for the antenna design in

(a) Photo of single antenna array. (b) Photo of LTCC RF front-end.

Figure 2.21: A 79 GHz LTCC radar front-end with 45-degree polarization antenna: (a) photo of single array and (b) photo of front-end. c 2015 IEEE [67].

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(a) LTCC antenna 3-D view and layer stack. (b) Photo of device: back view and front view.

Figure 2.22: Wide bandwidth LTCC radar front-end: (a) antenna structure and (b) photo of front-end. c 2017 IEEE [68]

LTCC material [67] (see Figure 2.21). Sickinger presented a very compact solution for the RF front-end of a 79 GHz radar system [68]. The total front-end size is only 2 cm by 3 cm. The antenna has a very wide bandwidth (>5 GHz) (see Figure2.22).

The multilayer material facilitates for complex feeding network in antenna designs, but it is not a full packaged solution for high-integrated antenna since the on-chip-off-chip connections are from either the bonding wire or the additional packaging.

The improvement in packaging technology brings new chance for the AiP solution. One of the earliest solutions is AiP based on quad-flat no-lead (QFN) package. The first QFN-based AiP was proposed by Gaucher in 2004 [69]. Prof. Zwick gave an insightful summary for QFN AiP development [70]. Figure2.23(a)shows a classic QFN package outlook, and Figure2.23(b)

shows an off-chip AiP configuration by QFN package.

Since another package technology—embedded wafer-level ball grid array (eWLB) —is avail-able for mmW applications [71, 72], a variety of AiP technologies in mmW applications have been developed [73]. Figure2.24shows some examples of AiP developed by eWLB for 77 GHz radar applications. Compared with a QFN package, eWLB eliminates the bonding wire in the packaging connections.

2.6 Examples of antenna in realized systems

This section gives a couple of examples of mmW antenna in industry products and gives a short comparison of them. Figure2.25shows antennas in four different automotive mmW radar products: middle-range radar from Bosch, long-range radar from Continental and short-range radars from Autoliv and Delphi, respectively.

The antenna in MRR of Bosch is a series-fed patch array structure. There are two Tx antennas. One is a high-gain antenna with maximum beam at broadside direction. The other Tx antenna is designed for tilted beam from the broadside direction. It is for elevation information evaluation. The distances between Rx antennas have nonuniform space. It is for better angle resolution.

The antenna in Continental LRR is a series-fed wire (open-ended transmission line) antenna. 21

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(a) The initial concept drawing for an antenna on top of chip within a QFN package [69].

(b) The concept of an off-chip antenna and wire-bond interconnect within a QFN package [74].

Figure 2.23: QFN AiP proposal of antenna on top of chip (a) and an off-chip interconnection drawing (b). c 2017 IEEE [70].

(a) AiP array at 60 GHz [75]. (b) AiP MIMO array at 77 GHz [76].

Figure 2.24: eWLB-based AiP solution of antenna array at 60 GHz (a) and MIMO array at 77 GHz (b). c 2018 IEEE [73].

There are some dummy antennas placed beside the Tx and Rx antennas. The purpose of the dummy antenna is to reduce the surface wave propagation inside the substrate.

In Autoliv SRR radar, there are three Tx antennas and four Rx antennas. The virtual antenna array concept is implemented for large equivalent aperture. All of those antennas are same series-fed patch array. It largely reduces the design procedure of the antenna. The separation between Tx channels has two different distances. The larger spacing is for better antenna resolution. The smaller spacing supports phase correction among different Tx channels. Delphi SRR has a complex structure of RF front-end. It separates MMICs and antennas on different sides of PCBs. The antenna structure is SIW antenna. This configuration reduces the parasitic radiation from the feeding structure, etc. Meanwhile, it also increases the system complexity and cost.

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(a) Middle-range radar from Bosch [77]. (b) Long-range radar from Continental [78].

(c) Short-range radar from Autoliv [79]. (d) Short-range radar from Delphi [80].

Figure 2.25: Examples of antenna in automotive radar from: (a) Bosch, (b) Continental, (c) Autoliv and (d) Delphi.

All of these antennas are selected planar structures because of their low profile and low cost in manufacturing. Two out of four antennas are patch antennas. The other two are wire antenna and SIW antenna, respectively. The RF substrate in most of the automotive radars is soft substrate like Rogers or Taconic laminate.

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Chapter 3 Differential Microstrip Patch

Antenna

3.1 Microstrip antenna

3.1.1 Microstrip structure

Microstrip line structure was first proposed by Grieg in 1952 [81]. Since then, it has become perhaps the most popularly used transmission line for radio frequency (RF) and microwave integrated circuits (ICs). This popularity and widespread use are because of its planar nature, ease of fabrication using photolithographic processes, easy integration with solid-state devices, easy combination with heat sink, good mechanical support and vast design information [82], [83].

The geometry of a microstrip structure is shown below in Fig3.1. A patterned conductor is printed on thin, fully grounded dielectric substrate of thickness h and relative permittivity εr. The wave traveling on microstrip is of the quasi-TEM mode. Figure3.2[84] shows a sketch of the field diagrams for the static approximation. The parallel-plate capacitor field dominates Ez, while at the conductor edges, the fringing fields dominate Ez and Ex. In the higher operating frequency, both E-field and H-field have small longitudinal components Ey and Hy.

Figure 3.1: Cross section of microstrip line structure.

3.1.2 Microstrip patch antenna

Microstrip structures were popular in circuit designs and, later on, were also used in antenna design. Microstrip patch antennas are planar antennas which are constructed under microstrip 24

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Figure 3.2: Electric and magnetic field lines at low frequencies with static approximation.

structures.

The first microstrip antenna concept was proposed by Deschamps in 1953 [85]. Microstrip antennas inherit the merit of microstrip structure and easy integration with planar structure circuits. However, it took 20 years before the first practical antennas were developed by Howell [86] and Munson [87]. A microstrip antenna in its simplest configuration consists of a radiating patch on one side of the dielectric substrate and a ground plane on the other side [88]. Ideally, the dielectric constant, εr, of the substrate should be low (2 < εr < 4) to enhance the fringe fields that account for the radiation.

Figure 3.3 illustrates a basic configuration of microstrip patch antenna. It consists of a very thin metallic strip (patch) placed a small fraction of a wavelength above a ground plane [89]. The radiating patch is a rectangular patch, fed by microstrip line. The thickness of the substrate is usually much less than the wavelength in the dielectric (h λr).

Figure 3.3: Microstrip patch antenna.

Microstrip antennas are referred to as patch antennas. The radiating patch may be square, rectangular, thin strip (dipole), circular, elliptical, triangular, or any other configuration. The radiating elements and the feed lines are usually photoetched on the dielectric substrate. The radiation mechanism of the patch can be considered as magnetic current (M) along the pe-riphery of the patch. The ground plane acts as mirror and will double the equivalent magnetic current of the patch.

Feeding methods of microstrip patch antenna can be categorized as line feed, coaxial feed, 25

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3. DIFFERENTIAL MICROSTRIP PATCH ANTENNA aperture-coupled feed and proximity-coupled feed [89].

Microstrip patch antennas have several advantages compared with other microwave antennas [88]. Some of the principal advantages of microstrip antennas are as follows:

• Lightweight, low-volume and thin profile;

• Low fabrication cost, suitable for mass production; • Linear and circular polarizations are possible;

• Dual-frequency and dual-polarization antennas can be easily made; • Fully ground structure;

• Easily integrated with microwave integrated circuits;

• Feed lines and matching networks can be fabricated simultaneously with the antenna structure.

Meanwhile, some disadvantages of microstrip antennas are as follows: • Narrow bandwidth and associated tolerance problems;

• Somewhat-lower gain;

• Large ohmic loss in the feed structure of arrays; • Most microstrip antennas radiate into half-space;

• Complex feed structures required for high-performance arrays; • Polarization purity is difficult to achieve;

• Poor end-fire radiator, except tapered slot antennas; • Extraneous radiation from feeds and junctions; • Lower power handling capability (-100 W); • Excitation of surface waves;

• Microstrip antennas fabricated on a substrate with a high dielectric constant are strongly preferred for easy integration with MMICs RF front-end circuitry. However, use of high dielectric constant substrate leads to poor efficiency and narrow bandwidth.

In particular mmW applications, for instance, a 77 GHz adaptive cruise control (ACC) radar, some of the disadvantages are minimized:

• Radiation into half-space is suitable for automotive radar applications;

• Few percent relative bandwidth is sufficient for the 76–81 GHz radar applications; • Broadside radiation is desired instead of end-fire radiation in the realized systems; • Power handling capability is limited with tens milliwatt.

Some of the other disadvantages may be improved by system designs like as follows: • Use differential feed mechanism to improve polarization purity;

• Improve antenna aperture by implementation of MIMO antenna configurations; • Improve RF front-end integration level with differential interface MMICs.

Therefore, microstrip patch antenna (MPA) is one of the most popular antennas in millimeter-wave radar applications.

3.1.3 Cavity model

There are many methods to analyze microstrip patch antenna, such as transmission line model, multiport network model and cavity model. The cavity model method gives physical insight and accurate results for microstrip patch antenna. It was first introduced by Lo and Richards in 1979 [90, 91]. This section gives a brief discussion of the cavity model and shows the fundamental results of the analysis.

There are three assumptions for the cavity model based on observation of the MPA on thin substrates (h λr). The following derivation is according to [92].

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The fields in the interior region do not vary with z (that is, ∂/∂z ≡ 0) because the substrate is very thin (h λr);

The electric field is z directed only, and the magnetic field only has the transverse compo-nents Hxand Hyin the region bounded by the patch metallization and the ground plane. This observation provides for the electric walls at the top and bottom;

The electric current in the patch normal to the edge of patch metallization is zero, which implies that the tangential component of ~H along the patch periphery is negligible, and a magnetic wall can be placed there. Mathematically, ∂Ez/∂n = 0.

The field distribution in the patch can be divided into two regions: the interior fields and the exterior fields. The interior fields are useful in determining the input impedance of the antenna and the currents responsible for radiation. The exterior fields are the fields outside the cavity region that determine the radiation characteristics of the patch antenna.

With fringing effects, the magnetic wall is placed at a distance ∆ away from the edges of the patch (see Figure3.4).

Figure 3.4: Magnetic wall of microstrip patch antenna.

Consider the region of the antenna between the patch metallization and the ground plane. Because the dielectric substrate is thin, the field distribution in this region can be described by TM to z modes with ∂/∂z ≡ 0. As a result, there are only three components of the fields Ez, Hxand Hy. The interior electric field ~Ei _{must satisfy the inhomogeneous wave equation.}

∇ × ∇ ~Ei− k2E~i≡ −jωµ0J~ (3.1) or ∂2Ez ∂X2 + ∂2Ez ∂y2 + k 2_Ez_{= jωµ0J} _(3.2) where k2 _{= ω}2_µ

0ε0εr, ~J is the excitation electric current density caused by either due to the coaxial feed or the microstrip feed, ˆz is a unit vector normal to the plane of the patch, and ∇ is the transverse del operator with respect to the z axis.

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3. DIFFERENTIAL MICROSTRIP PATCH ANTENNA

In addition to satisfying the wave equation, the fields must also satisfy the following bound-ary conditions:

ˆ

n × ~Ei_{= ˆ}_{n × ~}_Ee _{on the top and bottom conductors} _(3.3)

and ˆ n × ~Ei_{= ˆ}_{n × ~}_Ee ˆ n × ~Hi_{= ˆ}_{n × ~}_He ) on the walls. (3.4)

Here, ˆn is the unit outward normal to the walls, ~Ei _{and ~}_Hi _{are the fields in the interior} region, and ~Ee_{and ~}_He _{are the fields in the exterior region.}

Under the magnetic wall assumption, Equation3.4reduces to

ˆ

n × ~H = 0 on the magnetic walls (3.5)

It is now easy to determine the interior fields.

The electric field in the patch cavity can be written as

Ez= jωµ0I0 ∞ X m=0 ∞ X n=0 φmn(x1, y1)φmn(x0, y0)j2 0( mπd 2a ) k2_{− k}2 mn (3.6) ~ H = 1 jωµ0z × ∇Eˆ z (3.7)

where ω is the angular frequency and µ0 is the permeability of vacuum.

φmn(x, y) = pε0mε0n/aebecos(mπx/ae) cos(nπy/be) (3.8a) kmn2 = (mπ/ae) 2 + (nπ/be)2 (3.8b) k2 = k₀2εr(1 − jtanδ) (3.8c) k0 = ω/c = 2πf /c (3.8d) j0(x) = sin(x)/x (3.8e)

The magnetic current (M ) around the periphery of the patch may be calculated as

M = −2ˆn × Ez (3.9)

Figure 3.5 shows the first several modes of MPA, TM01, TM10, TM20, TM21, etc. When the feeding point is located at (a/2,0), the fundamental is TM01. When the feeding point is located at (0, b/2), the fundamental is TM10.

Since the electric current on patch is negligible, the radiation performance can be calculated from M . The uniform M distribution edges represent radiation edge since they are main 28

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contributors to the radiation of the patch. The nonuniform M distribution edges are non-radiation edges. Here, non-non-radiation edges mean those edges that have very little non-radiation on the principle plane. The higher-order modes introduce feed reactance and cross-polarization radiation of the patch [89].

(a) TM01mode (b) TM10mode

(c) TM20mode (d) TM21mode

Figure 3.5: Electric field and magnetic surface current distributions along the periphery for various modes of a rectangular microstrip antenna.

The input impedance can be calculated as

Zin= Vin

I0 (3.10)

where Vinis the RF voltage at the feed point. It is computed from Equation3.6as

Vin = −Ez(x0, y0)h (3.11)

= −jωµ0hI0 ∞ X m=0 ∞ X n=0 φ2_mn(x0, y0)j₀2(mπd 2a ) k2 mn− ke2 (3.12) 29

Differential feed antenna in millimeter wave Radar applications / submitted by Ziqiang Tong

Differential feed antenna

in millimeter wave Radar

applications

Doctoral Thesis

to obtain the academic degree of

Doktor der technischen Wissenschaften

in the Doctoral Program

Acknowledgements

Abstract

Contents

List of Figures

Chapter 1

Introduction

1.1

Radar

1.1.1

FMCW radar

1.1.2

MIMO radar

1.1.3

Radar in millimeter-wave applications

1.2

Motivation

1.3

Simulation tool

1.4

Thesis structure

Chapter 2

State of Art for mmW Radar

Antenna

2.1

Waveguide antenna

2.2

Lens antenna

2.3

Reflector antenna

2.4

Planar antenna

2.4.1

Open-ended transmission line antenna

2.4.2

Grid antenna

2.4.3

Patch antenna

2.4.4

Substrate integrated waveguide (SIW) antenna

2.5

High-integration antenna

2.5.1

Antenna on chip

2.5.2

Antenna in package

2.6

Examples of antenna in realized systems

Chapter 3

Differential Microstrip Patch

Antenna

3.1

Microstrip antenna

3.1.1

Microstrip structure

3.1.2

Microstrip patch antenna

3.1.3

Cavity model