mmW-DMPA design at 79 GHz

3. DIFFERENTIAL MICROSTRIP PATCH ANTENNA

Figure3.14shows the real part of the antenna impedance of DMPA with edge feed and middle feed. It shows that (i) the edge feed MPA has more modes than middle feed, for instance, TM11, TM₃₁, etc., and that (ii) around the fundamental mode TM₀₁, both DMPAs have a similar real part of antenna impedance Real(Z).

Figure3.15shows the imaginary part of the antenna impedance of DMPA with edge feed and middle feed. It shows that around the fundamental mode TM01, the Imag(Z) of edge feed is higher than that of middle feed DMPA. The reason is that the higher-order mode TM11 raises the Imag(Z).

Table3.1shows the simulated results of the patch. For the same patch, the edge feed requires lessy_xfor the impedance matching. That is mainly because of the influence of the higher-order mode TM11.

The theory of edge-feed analysis has been published by the author in [112].

Table 3.1: Simulated DMPA with middle feed and edge feed

Port type Feed method W L y_x

lumped middle feed 1.30 mm 0.98 mm 0.3 mm lumped edge feed 1.30 mm 0.98 mm 0.2 mm

Figure 3.13: Real(Z) of DMPA with edge feed.

Figure 3.14: Real(Z_{DM P A}) of DMPA middle and edge feed.

Figure 3.15: Imag(Z_{DM P A}) of DMPA middle and edge feed.

Two feeding configurations, middle feed and edge feed, are further discussed with coupled microstrip feeding lines [107,113] (see Figure3.16).

Figure 3.16(a) shows the middle feed configuration. The coupled MSLs connect to the radiating edges – the top and bottom side of the patch. It uses an inset to adjust the feed positions on the patch. The advantage of the middle feed configuration is less high-order mode resonance. But it has length and bending feeding lines, which introduce loss and radiation in system integration. In addition, the inset destructs the radiating edge of the patch.

Figure3.16(b) shows the edge feed method. The coupled MSLs connect to the patch from the non-radiating edge – the left side of the patch. The impedance matching can be realized by adjusting the geometry of the couple microstrip lines (WmandSm). It has a compact and straight feeding line structure in system integration. The radiating edges of patch are kept completely. Furthermore, edge feeding is more feasible for use in an antenna array. Therefore, in this work, edge feeding structures are proposed for DMPA design.

(a) middle feed (b) edge feed

Figure 3.16: DMPA with (a) middle feed MSL and (b) edge feed MSL.

The initial DMPA model as in Figure 3.16(b) is built in CST MWS. The patch sizes are inherited from the previous section, where Lp = 0.98 mm and Wp = 1.3 mm. The feeding points in the previous sections are replaced by coupled MSLs.

Since the feeding line is DMSL, it is necessary to give an analysis of DMSL. Figure 3.17 shows the structure of coupled MSLs. It has two modes of signals: common mode signals and differential mode signals. The characteristic impedance of the coupled MSLs also has different 41

3. DIFFERENTIAL MICROSTRIP PATCH ANTENNA

values under the different modes [114]. For a fixed substrate (dielectric constantrand thickness h), coupled MSLs have two parameters (Smand Wm) to adjust the characteristic impedance, while single-ended MSL has only one parameter (W_m). This greatly assists the impedance matching of the DMPA.

Figure 3.18shows the simulated differential mode characteristic impedance of the coupled MSL at 79 GHz. There are multiple pairs of (Sm, Wm) having the same Zd as 100 Ohm, for instance, (Sm, Wm) = (225 µm, 260 µm), or (325 µm, 280 µm), etc. When coupled MSLs connect to the patch, different pairs correspond to different feeding positions of the patch. Following the example in Table3.1, it is straightforward to select the initial geometry of coupled MSLs as feeding lines. Here, we consider that the feeding point of coupled MSLs has the middle of the lines. The initial feeding point is selected asy1= 0.2 mm, which corresponds toWm+Sm=L−2∗y1= 0.58 mm.

Till now, we set up a simulation model of DMPA with coupled MSLs edge feed. The model has been built and simulated in CST MWS. The simulation port is a rectangular waveguide port, which is more accurate for port de-embedding in the simulation. CST MWS supports the setup of high-order modes for the waveguide port. In this way, both common mode and differential mode behaviors are simulated.

Figure 3.19shows the simulated reflection coefficient of the initial model, both differential mode and common mode. The differential mode reflection coefficient shows the resonance around 80 GHz, which corresponds to the resonance mode TM01. For the common mode signal, the resonant frequency is on TM10and TM20mode, which is below 65 GHz and above 120 GHz, respectively. In the desired frequency range (76-81 GHz), the common mode signal behaves as open-ended with the chosen length (Wp) of transition line. Therefore, the antenna has good differential-to-common mode rejection ratio.

Figure 3.17: Coupled MSL structure.

Figure 3.20shows the E-field distribution of differential mode signal at 79 GHz. It shows uniform distribution along x-axis and nonuniform distribution along y-axis. It proves the fun-damental mode of the patch is TM₀₁mode.

The parameters of the single patch DMPA are studied accordingly. Figure 3.21 shows that varying L_p corresponds to a resonant frequency change; an increase in L_p will decrease the resonant frequency fr. Figure 3.22 shows that the feeding distancey1 shifts the antenna impedance from high Ohmic range to low Ohmic range.

Figure3.23shows the relative bandwidth of DMPA for theW_p/L_pratio varying from 0.7 to 1.6. It shows that the maximum relative bandwidth occurs when the ratio ofWp/Lp is between 1.3 and 1.4. Further comparison between the proposed DMPA and SMPA fed by MSL are also 42

Figure 3.18: Simulated differential mode characteristic impedance of coupled MSL at 79 GHz.

Figure 3.19: Single patch DMPA simulated return loss (RL) in Smith Chart.

shown here. Compared with SMPA, DMPA shows wider relative bandwidth whenW_p/L_p<1.5.

The maximum BW% reaches 4.8% for DMPA, while it is 4.55% for SMPA. The major reason is that the inset of the patch degrades the bandwidth of SMPA. When Wp/Lp > 1.5, the TM11mode starts to degrade DMPA bandwidth. DMPA and SMPA show the same BW when Wp/Lp= 1.6.

The DMPA has been further optimized for center frequency at 79 GHz and bandwidth. The S-parameters of the final optimized patch are shown in Figure 3.24. In the desired frequency range, the single patch DMPA shows 3.3 GHz bandwidth for 10 dB return loss of differential signal, while the reflection for common mode (CM) signal is very high (below 1 dB return loss).

During the optimization procedure, some design rules are concluded. For instance, 1) Lp controls the differential mode resonant frequency (TM01) of the patch;

2) y_i controls the impedance match; and

3) Wp controls the relative bandwidth, as well as the common mode resonant frequency.

43

3. DIFFERENTIAL MICROSTRIP PATCH ANTENNA

Figure 3.20: Simulated differential mode E-field distribution at 79 GHz.

Figure 3.21: Parameter study of single patch DMPA –Lp.

The designed single patch DMPA has been manufactured and measured. The RF material is selected as RO3003 with 127µm thickness. The first step is the S-parameter measurement.

Figure3.25shows the prototype DMPA for S-parameter measurement. Since it is not easy to have direct measurement of the differential signal response, the DMPA was first measured as a two-port device. A taper structure was built between coupled MSLs and the ground-signal-ground (GSG) probe stub (see Figure3.25). The vector network analyzer(VNA) and frequency extender were used as the measurement devices. The S-parameters as well as the radiation patterns of the prototype can then be measured.

The antenna was measured as a two-port device, and the reflection coefficient was calculated by mixed-mode matrix [110].

44

Figure 3.22: Parameter study of single patch DMPA –y₁.

Figure 3.23: Relative bandwidth of DMPA and SMPA with differentWp/Lpratio.

Figure 3.24: Simulated reflection coefficient for the optimized DMPA with size Lp = 0.98mm, Wp= 1.34mm, Wm= 0.26mm, Sm= 0.20mm.

45

3. DIFFERENTIAL MICROSTRIP PATCH ANTENNA

M = 1

√2

"

1 −1

1 1

#

(3.44)

SM M = M ·SSE·M⁻¹ (3.45)

with SSE as the measured single-ended two-port S-parameters andSM M as the calculated one-port differential S-parameters.

Figure3.26shows the measurements in comparison with the simulation results of the reflec-tion coefficient of the two antennas. The measured bandwidth of a single patch DMPA is 4.7 GHz. It is wider than the simulation results because of the losses in the substrates. Meanwhile, the common mode return loss shows high reflection from 60 GHz to 90 GHz.

The next step is to measure the radiation pattern of the single patch DMPA. In a mmW measurement setup, coaxial cable is expensive and difficult to use to fulfill the measurement setup. Active devices with waveguide interface, like harmonic mixer, are more popularly im-plemented. Therefore, it is convenient to build a transition which can convert signal from the rectangular waveguide to the symmetric MSL.

Figure 3.27 shows the prototype of DMPA for far-field measurement. It is composed of a transition from a waveguide to coupled MSLs and a single patch DMPA. The transition converts signal from rectangular waveguide to the differential mode signal on coupled MSLs.

The DMPA prototype was measured as a receive antenna. Section 3.4gives further details of the transitions.

The normalized E-plane and H-plane, co-polarization and cross-polarization radiation pat-terns of the single patch DMPA are plotted in Figures3.28and3.29. The measurement shows that the half power beamwidth (HPBW) of single patch DMPA is 88 degrees in E-plane and 62 degrees in H-plane, respectively. The simulation results of the cross-polarization in H-plane are neglected here because of its quite low level.

Additional noise is observed within the far-field measurements. Therefore, average values (20 samples per degree) were used for plotting the results. Because of the height of the transition cap, the last several degrees of the radiation patterns in the H-plane [85 to 90 degrees] are disturbed.

Within the far-field measurements, some ripples were observed in the radiation pattern, stronger in the E-plane than in the H-plane. Radiation from the surface wave at the edges of PCBs could be the most likely explanation. It is a well-known fact for a microstrip type of antenna [115]. Therefore, an absorbing material was used on the edges of PCBs. This reduced the surface wave effects but also distorted the radiation patterns at the angle ranges [+(-)80 to +(-)90 degrees].

The gain of single patch DMPA was measured using a comparison with a standard horn antenna. The calibrated gain for the single patch DMPA is 6.2 dBi at 79 GHz.

From the measurement results of the prototype, the concept of single patch DMPA was approved. The results were first reported by the author in [113].

46

Figure 3.25: Single patch DMPA prototype for S-parameter measurement.

Figure 3.26: Simulated and measured reflection coefficient of DMPA.