Improvement for common-mode signal suppression:

Prototype design in W-band

In the previous section, the gap-coupled transition has a narrow band resonance at 79 GHz for the common-mode signal. Further study shows that the common-mode resonance is coming from the parasitic patch. When the concept extends to W-band, the common-mode signal rejec-tion is also worse. The return loss of common-mode signals may reach 3 dB. Such common-mode signals within the transition either leak to the substrate or radiate from the waveguide port.

To suppress the common-mode signal, a new design with center frequency at 94 GHz is shown in this section. In addition, the matching network design is more robust for manufacturing.

The structure of such a transition is shown in Figure 4.24. It has a similar structure as before, while the waveguide is selected WR10 for W-band applications. The main change is the DMPA part. In this work, one side of the parasitic patch is connected to the ground to 83

4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION

Figure 4.23: Measurement results of GCP transition on RO3003 and TLE95 material.

facilitate the matching network design and suppress the common-mode signals. The substrate material of the PCB is Taconic TLE-95 (ε_r=2.95,h=0.127 mm, tanδ=0.004). The conductivity of the coupled MSLs is 5.8e7 S/m. The transition housing has a waveguide interface for WR10 and a channel for coupled MSLs.

The transition housing is a modified rectangular waveguide section (see Figure 4.24). A channel which allows signal propagation along the coupled MSLs is cut into the short side wall of the waveguide. The dimension of the channel (H_c and W_c) should not be so small as to interfere with the wave propagation on the PCB. However, ifHc is too high, it causes waves to leak from the transition. In this work,H_cwas chosen to be 1.0 mm, which is about eight times the thickness of the substrateh. Wc was chosen to be 5.0 mm.

Figure 4.24: Structure of vertical transition between rectangular waveguide and coupled MSLs – short-ended parasitic patch. c2012 IEEE [148].

The top view of the PCB is shown in Figure 4.25. The antenna includes the radiation part (the main patch and the parasitic patch) and the matching network. The main patch of the antenna is fed by coupled MSLs. The parasitic patch is side placed to the main patch. The electromagnetic power is coupled through the non-radiating edge of both patches. Different 84

from the previous design, the other side of the parasitic patch – the right side of the patch in the figure – is connected to the ground by vias. This modification brings two advantages in the design: (1) it suppresses the common-mode signal in the center frequency, and (2) it facilitates the matching network design.

Figure 4.25: Top view of the PCB of the transition – W-band transition with common mode suppression. c2012 IEEE [148].

There are two kinds of propagation modes in the coupled MSLs: the differential mode and the common mode. Figure 4.26shows the simulated E-field distribution of the antenna under both mode of input signals. The radiating edges of the patches, which have uniform E-field dis-tribution, are parallel to the long side walls of the waveguide (x-axis) for the differential-mode feeding signals, whereas they are parallel to the short side walls (y-axis) for the common-mode feeding signals. Consequently, the fundamental common-modes of the patch are TM01 mode for differential-mode signals and TM₁₀ mode for common-mode signals. TM₀₁ mode also matches with the fundamental mode of the waveguide. The match of signal modes allows the transmis-sion of energy.

In the case of differential signals, the resonant frequency of the TM01 mode on each patch is determined by the patch length (l₃, l₄). It is slightly less than λ_r/2 because of fringing effects. Here,λris the wavelength in the dielectric layer. Dual resonant frequencies are realized by setting l3 and l4 to different values. The patch widths (w3, w4) are the main factors in determining the bandwidth of the transition. Therefore, for the main patch, w3 is slightly larger thanl3. For the parasitic patch, because of the short-end edge,w4 is much longer than l₄to support the desired TM₀₁ mode.

In the case of common-mode signals, the resonant frequency of the TM10mode on the patch is determined by the patch lengths (w₃,w₄), and because of the feeding lines of the main patch and the short-ended edge of the parasitic patch, the TM10 modes are suppressed in the desired frequency range. Butw₄ should not be longer than 1.5λ_r; otherwise, higher order modes arise.

The patch impedance (in the A-A’-plane) has an inductive part. Therefore, two sections of coupled MSLs are built for the impedance match. The matching network is similar to the previous design. The first one, from A-A’-plane to B-B’-plane, is shorter than a quarter of the wavelength. It converts the patch impedance into a value greater than 100 Ohm on 85

4. COUPLED MICROSTRIP LINE FEED WAVEGUIDE TRANSITION

the real axis of the Smith Chart. The second one, from B-B’-plane to C-C’-plane, has a high characteristic impedance. It works as a quarter-wavelength transformer and matches the 100 Ohm differential-mode impedance of coupled MSLs. If the inductive part of the patch impedance (at the A-A’-plane) is too high, the matching network is difficult to realize because of fabrication limitations. The short-ended parasitic patch reduces the inductive value of the patch impedance, which facilitates realization of the matching network.

(a) E-field of the differential-mode signal at 93 GHz

(b) E-field of the differential-mode signal at 100 GHz

(c) E-field of the common-mode signal at 96 GHz

Figure 4.26: E-field distribution of transitions at differential mode (a-b) and at common mode (c). c2012 IEEE [148].

After optimization with CST MWS, a prototype of design is achieved. The S-parameter results of the transition with the optimized dimensions are shown in Figure4.27. The simulation

Figure 4.27: Simulated S-parameters of the transition (l1=0.53 mm, w1=0.13 mm, S1=0.50 mm, l2=w2=0.25 mm, S2=0.26 mm, l3=0.81 mm, w3=0.92 mm, l4=0.78 mm, w4=1.05 mm, g=0.11 mm, d=0.4 mm, D=0.25 mm, a=2.54 mm, b=1.27 mm, h=0.127 mm, Hc=1.0 mm, Wc=5.0,εr=2.95). c2012 IEEE [148].

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Figure 4.28: Photo of W-band transition back-to-back structure with shim. c2012 IEEE [148].

model is also shown at the left corner. Both ports are set as rectangular waveguide port in CST MWS. From the simulation results, the 10 dB return loss bandwidth of differential mode signals is about 15.9 GHz (BW% = 16.6%). The return loss of the waveguide port exhibits a similar behavior. The insertion loss of the differential mode signals to waveguide is 0.4 dB at the center frequency (96 GHz). The common-mode return loss of the coupled MSLs port is only 0.5 dB within the bandwidth and less than 1 dB for the whole W-band. The insertion loss of the common mode to waveguide (or to differential mode) is too low to plot.

The back-to-back (B2B) structures of the designed transition with different connecting line lengths were fabricated and subsequently measured. For ease of fabrication, the transition housing was simplified into a 1 mm thick transition slice – shim (see Figure 4.28). Connecting it to a standard WR10 waveguide flange results in the transition housing depicted in Figure 4.24. It is a cheaper solution compared with a full-metal housing as in Figure4.14.

Figure 4.29: Measurement and simulation results of B2B structure of the transition with dif-ferent lengths of the connecting lines, 21 mm and 31 mm, respectively. c2012 IEEE [148].

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The measurement and the simulation results of the B2B structure are plotted in Figure4.29.

The bandwidth is 14.5 GHz for 10 dB return loss and 10.8 GHz for 15 dB return loss. It matches the simulation results. The frequency shift is caused by fabrication tolerances. The insertion losses were 3.7 dB and 5.0 dB at 96 GHz for short (21 mm) and long (31 mm) connecting lines, respectively. Therefore, the insertion loss for a single transition is 0.5 dB. The loss tangent of the material (tanδ=0.004) quoted by Taconic is only up to 20 GHz. This introduces a difference between the measured and the simulated insertion loss. If loss tangent (tanδ) is taken as the dominant factor, neglecting the surface roughness of conductors, a value of tanδ=0.008 fits the measurement well.

Repeatability Test

The repeatability of the test boards were tested in two different ways. The measurement results are shown in Figure4.30. In the first test, the top mount part was dismounted and reassembled five times (see Figure 4.30(a)). It verifies the mounting repeatability on the same board. In the second test, three PCB samples with the same design were measured to verify the PCB manufacturing stability (see Figure4.30(b)). In both tests, the measurement results show very good repeatability. It proves the transition design is a robust solution. It also shows the simple solution of shim supports good repeatability.

(a) Repeated measurements of the same board of B2B structures of the transitions.

(b) Measurements of the different boards of B2B structures of the transitions.

Figure 4.30: Repeatability test of reassembling one same board (a) and different PCB samples (b).