SiGe-Based Broadband Integrated Circuits and Systems for Millimeter- Wave & THz Radar Applications / submitted by Faisal Ahmed

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Submitted by Faisal Ahmed Submitted at

Institute for Communi-cations Engineering and RF-Systems Supervisor and First Examiner Univ.-Prof. Dipl.-Ing. Dr. Andreas Stelzer Second Examiner Univ.-Prof. Dipl.-Ing. Dr. Nils Pohl August 2018 JOHANNES KEPLER UNIVERSITY LINZ Altenbergerstraße 69 4040 Linz, ¨Osterreich www.jku.at DVR 0093696

SiGe-Based Broadband Integrated

Circuits & Systems for

Millimeter-Wave & THz Radar Applications

Doctoral Thesis

to obtain the academic degree of

Doktor der technischen Wissenschaften

in the Doctoral Program

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Dissertation

SiGe-Based Broadband

Integrated Circuits and Systems

for Millimeter-Wave & THz

Radar Applications

Faisal Ahmed

Institute for Communications Engineering and RF-Systems

Johannes Kepler University, Linz, Austria

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Faisal Ahmed: SiGe-Based Broadband Integrated Circuits and Systems for Millimeter-Wave &

THz Radar Applications, © August 2018.

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Statutory Declaration

I hereby declare that the thesis submitted is my own unaided work, that I have not used other than the sources indicated, and that all direct and indirect sources are acknowledged as references.

This printed thesis is identical with the electronic version submitted.

Linz, ___________ ______________________

Date Signature

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Acknowledgments

Foremost, I would like to express my sincere gratitude to Univ. Prof. Dipl.-Ing. Dr. Andreas Stelzer for giving me an opportunity to be a part of his wonderful team at the Institute for Communications Engineering and RF-Systems. I feel privileged to learn, work, and collaborate with him over these past years. His support, guidance and supervision throughout the phase of this work has been exemplary. I thank Univ. Prof. Dipl.-Ing. Dr. Nils Pohl from Ruhr-Universität Bochum for being my 2nd examiner. I am also deeply thankful to Univ. Prof. Dipl.-Ing. Dr. Andreas Springer for his kind and gentle support, be it technical or administrative. I would like to acknowledge the funding support from the Austrian Center of Competence in Mechatronics, the EU-funded DOTSEVEN project, and ECSEL-JU TARANTO. I would also like to express my gratitude to Assist. Prof. Dipl.-Ing Reinhard Feger for sharing his knowledge and experience in FMCW radars systems. Dr. Martin Jahn was an influential figure during my early days in the world of high frequency integrated circuit design and I am grateful to his valuable guidance. I am also very thankful to Dr. Abouzar Hamidipour, Dr. Herman Ng, and Dr. Kambiz Hadipour for numerous technical discussions, inspirations, and brainstorming sessions. The experimental demonstration of this work would not have been possible without the support of ADir Ralf Rudersdorfer and Richard Hütner. The work presented in this thesis is directly focussed on SiGe HBT technology, and I was lucky enough to have learnt from none other than the top technology experts Dr. Klaus Aufinger, Dr. Walter Hartner from Infineon Technologies and Dr. Bernd Heinemann from IHP Microelectronics. During this Ph.D. I was fortunate to have many admirable colleagues with whom I shared the office and had a wonderful time. I thank them all: Dr. Xin Wang, Dr. Domink Zankl, Tong Ziqiang, M.Sc, and specially Matthias Wakolbinger, M.Sc. I would like to mention in this list my institute’s colleagues: Dr. Christian Schmid, Dipl-Ing. Thomas Wagner, Dipl.-Ing. Heinz Haderer, Dipl.-Ing. Sebastian Poltschak, Dipl.-Ing Werner Scheibelhofer, and Matthias Porranzl, M.Sc. I would also like to specially thank Monika Scheuchenegger for her continuous administrative support. I am grateful to my current colleagues at DICE Dr. Christoph Wagner and Dipl.-Ing Karl Dominizi in supporting me in the final stages of my thesis. A very special thanks to my good friend and colleague Muhammad Furqan, M.Sc. for his enduring support and without whom this entire journey would have been much less enjoyable and successful. Most importantly, I cannot express enough gratitude to my inspirational father, my loving mother, my caring brother, and my sweet sister. And last but not the least, I will always be indebted to my wife for her unconditional support, encouragement, and unfaltering love.

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Abstract

Today’s society is set on its course for multi gigabits-per-second wireless connectivity, internet of things, non-invasive medical diagnosis, intelligent mobility, and fully automated driving. This is a paradigm shift, one which requires constant advancements in the technologies present at the core of these applications. Furthermore, making these applications viable for mass-market requires regulating the cost and, depending upon the application, high reliability. Silicon-germanium (SiGe) heterojunction bipolar transistor (HBT) technology has proven itself as one of most valuable technology enabling systems operating at millimeter-wave (mm-wave) frequencies and beyond. Moving to higher frequencies results in increased absolute bandwidth, but based on past trends and future forecasts, there is a continuous demand on increased bandwidth which coarsely corresponds to higher information rate in communications or higher resolution in radars. This bandwidth requirements at the moment can not be simply met using conventional architectures and calls for intensive research. The work presented in this thesis is motivated by these factors and focusses on achieving very high fractional bandwidths in circuits operating at mm-wave and sub-millimeter wave frequencies. Although, some of the circuits presented here can be used in a wide variety of application scenarios, mostly are adapted for high resolution radar sensors.

Two advanced 130 nm SiGe BiCMOS technolgies, one from Infineon Technogies and the other from IHP Microelectronics, have been used for this work. Starting with the design of broadband amplifiers, compact lumped circuit based amplifiers with a bandwidth extending from DC to 105 GHz are presented. These lumped circuit amplifiers are based on a novel topology of cascaded common-base, emitter-followers, and an output cascode stage which improves the performance on several aspects as compared to similar lumped architectures present in the existing literature. A thorough analysis based on analytical expressions of voltage transfer characteristics and input impedance is provided which aids in comprehending the circuit dynamics and in the design process. Special focus is given on stability issues of emitter follower chains which are prone to produce ringing. Another amplifier with a 3-dB fractional bandwidth of more than 50% and a peak differential output power of 11 dBm is presented. It is the first Si-based amplifier which covers the entire D-band frequency range (110−170 GHz). Based on four interstage-matched cascode amplifiers, it uses a T-type four-reactance matching network together with optimized HBT dimensions to construct a uniform gain profile. The ix

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efficacy of this technique is explained in terms of analytically calculated transimpedance gain profiles of the interconnected stages. By using these profiles and gain staggering broadband amplifiers can be designed also in other frequency bands. Measuring differential broadband circuits using single-ended equipment is a challenge. Therefore, during the course of this work broadband Marchand baluns operating in the D-band and in the J-band (220−325 GHz) were designed based on a three-symmetric line modified topology which reduces the phase velocity difference between odd and even modes. Highly linear downconversion mixers with broadband conversion gain (CG) are essential for realizing high resolution radar sensors. In this work, a fundamental-wave and a novel subharmonic downconversion receivers are presented working in the D-band and J-band with a 3-dB CG bandwidth of 30 GHz and 73 GHz, respectively. An integral part of high performance radar sensors are high-power, low-noise signal sources based on either frequency multipliers or voltage controlled oscillators (VCOs). Design and analysis of these signal source working up to 300 GHz is a core part of this thesis. A fundamental-wave D-band signal source with a figure-of-merit of −185 dBc/Hz and an output power level of around 9 dBm around 160 GHz is presented. For signal sources operating in the J-band harmonic approaches must be utilized. By using a push-push topology the operating frequency can be doubled, however the resulting output power usually turns out to be much lower. In this work, a novel technique for improving the output power and efficiency of push-push based VCOs is proposed which in the presented case is able to enhance the output power of two 300 GHz VCOs by a factor of around 5 dB as compared to the conventional approaches. A complete theoretical analysis of the technique is provided by means of calculating the common-mode impedance of the VCOs.

Towards the end of this work a frequency-modulated continuous-wave (FMCW) radar sensor in an embedded-wafer-level-ball-grid-array package working in the J-band is designed relying on many of the aforementioned techniques. At these frequencies antenna-on-chip, for most of the cases, is the only viable solution. The gain of the system can then be subsequently increased either by using a lens or a waveguide horn antenna thereby making the system bulky and increasing the form factor. In this work, for the first time an antenna-in-package working at more than 240 GHz is designed and implemented. The antenna shows a measured gain of more than 4 dBi and a 3-dB gain bandwidth of 24 GHz.

The research carried out during this work demonstrates that the SiGe HBT technology shows a tremendous advantage in meeting not only the rising demands of current mm-wave systems but it also possesses a enormous capability for future THz applications. This however constantly requires adapting advanced and state-of-the-art circuit concepts and architectures so that the potential of this technology can be fully exploited.

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Kurzfassung

Die heutige Gesellschaft ist klar auf Kurs Richtung drahtloser Kommunikation mit mehreren GB/s, Internet der Dinge, nicht-invasiver medizinischer Diagnostik, intelligenter Mobilität und vollautonomen Fahren. Dies stellt einen grundsätzlichen Wandel dar, der eine stetige Weiteren-twicklung von vorhandenen Technologien, die den Kern der Anwendungen bilden, fordert. Um diese Anwendungen massentauglich zu machen sind außerdem eine Reduktion der Kosten und eine hohe Zuverlässigkeit erforderlich. Silizium Germanium (SiGe) Bipolartransistoren mit Heteroübergang haben sich als eine der nützlichsten Technologien herausgestellt um Systeme zu ermöglichen die bei Millimeterwellenfrequenzen und darüber operieren. Die Bewegung zu höheren Frequenzen resultiert in höherer absoluter Bandbreite. Der Trend der Vergangenheit und Prognosen über die Zukunft zeigen einen kontinuierlichen Bedarf an Bandbreite, welche zu höheren Datenraten in Kommunikationssystemen und höherer Auflösung bei Radaren führt. Die aktuellen Anforderungen an die Bandbreite können nicht einfach mit gewöhnlichen Architekturen realisiert werden, sondern fordern intensive Forschung. Die vorliegende Arbeit ist daraus motiviert und fokussiert sich auf die Realisierung von Schaltungen hoher Bandbreite bei Millimeterwellenfrequenzen und darüber. Obwohl einige der Schaltungen für unterschiedlichste Anwendungen eingesetzt werden können, sind sie primär auf hochauflösende Radarsensoren ausgerichtet.

Für die vorliegende Arbeit werden zwei fortgeschrittene 130 nm SiGe Technologien eingesetzt: eine von Infineon Technologies und die andere von IHP Microelectronics. Zunächst werden breitbandige Verstärker die von Gleichstrom bis 105 GHz funktionieren präsentiert. Diese LC-basierten Breitbandverstärker haben eine neuartige Topologie von kaskadierter Basisschaltung, Emitterfolgern und einer Ausgangskaskode, wobei die erzielten Ergebnisse Publikationen mit anderen Architekturen in einigen Aspekten übertreffen. Eine detaillierte Analyse, gestützt auf analytischen Ausdrücken der Spannungsübertragungsfunktion und der Eingangsimpedanz, ist angegeben und hilft beim Verständnis der Schaltungsdynamik und dem Design Prozess. Spezieller Fokus wird dabei auf Stabilitätsprobleme der Emitterfolger Kette, welche anfällig ist auf transiente Einschwingvorgänge, gelegt. Ein weiterer Verstärker mit einer relativen 3-dB-Bandbreite von über 50 % und einer differenziellen Ausgangsleistung von bis zu 11 dBm wird ebenso präsentiert. Es handelt sich um den ersten Silizium-basierten Verstärker der den gesamten D-Band Frequenzbereich (110−170 GHz) abdeckt. Aufbauend auf vier

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chenstufenangepassten Kaskodenverstärken, benutzt er eine Type T Anpassschaltung mit vier reaktiven Elementen gemeinsam mit optimierten HBT Geometrien um eine gleichmäßige Verstärkung zu realisieren. Die Wirkung dieser Technik wird über analytische Ausdrücke der Transimpedanzverstärkung der verschalteten Stufen erklärt. Mittels dieser Ausdrücke und gestaffelter Breitbandverstärker lassen sich so auch Schaltungen für andere Frequenzbänder realisieren. Differentielle Breitbandschaltungen mit unsymetrischem Messequipment zu charak-terisieren ist schwierig. Deshalb wurden in dieser Arbeit auch breitbandige Marchand Baluns für den Betrieb im D-Band und im J-Band (220−325 GHz) ausgelegt, basierend auf einer modifizierten Topologie mit drei symmetrischen Leitungen, welche die Phasengeschwindigkeits-differenz zwischen den geraden und ungeraden Moden reduzieren. Mischer mit hoher Linearität und breitbandiger übertragungsverstärkung sind essenziell für hochauflösende Radarsensoren. In dieser Arbeit werden ein Grundwellenmischer und ein subharmonischer Mischer vorgestellt, die im D-Band und im J-Band jeweils eine 3-dB-Bandbreite von 30 GHz bzw. 73 GHz aufweisen. Ein integraler Bestandteil von hochperformanten Radarsensoren sind Signalquellen mit hoher Ausgangsleistung und geringem Rauschen, basierend auf entweder Frequenzvervielfachern oder spannungsgesteuerten Oszillatoren (englisch voltage controlled oscillators, VCO). Die Ausle-gung und Analyse solcher Signalquellen bis zu 300 GHz ist ein zentraler Teil der vorliegenden Arbeit. Eine Grendwellen-Signalquelle für das D-band wird präsentiert, mit einer Kennzahl von -185 dBc/Hz und einer Ausgangsleistung von 9 dBm bei ca. 160 GHz. Für Signalquellen im J-band müssen harmonische Konzepte angewendet werden. Mit einer Gegentaktstufe kann die Ausgangsfrequenz verdoppelt werden, jedoch führt dies zu einer Einbuße der Ausgangsleistung. In dieser Arbeit wird eine neuartige Technik vorgestellt, welche die Ausgangsleistung und Effizienz von Gegentakt-VCOs verbessert. Im vorgestellten Fall wird die Ausgangsleistung von zwei 300 GHz VCOs um ca. 5 dB gegenüber klassischen Herangehensweisen verbessert. Eine umfassende theoretische Analyse der Technik basierend auf der Berechnung der Gleichtak-timpedanz der VCOs wird vorgestellt. Gegen Ende der Arbeit wird ein frequency-modulated continuous-wave (FMCW) Radar Sensor in einem embedded-wafer-level-ball-grid-array Gehäuse präsentiert, der im J-Band arbeitet und auf vielen der zuvor genannten Techniken basiert. Bei diesen hohen Frequenzen sind zumeist nur am Chip integrierte Antennen passable Lösungen. Die Richtwirkung solcher Systeme kann durch Linsen oder Hohlleiterantennen verbessert werden, jedoch führt dies zu unhandlichen Systemgrößen und Formfaktoren. In dieser Arbeit wird zum ersten Mal eine im Gehäuse integrierte Antenne bei mehr als 240 GHz ausgelegt und implementiert. Die resultierende Antenne zeigt eine gemessene Richtwirkung von mehr als 4 dBi und eine 3-dB Bandbreite von 24 GHz.

Die durchgeführte Forschungsarbeit zeigt, dass SiGe nicht nur große Vorteile in Bezug auf den steigenden Bedarf an Millimeterwellensystemen, sondern auch enorme Einsatzmöglichkeiten für zukünftige THz Anwendungen besitzt. Wenngleich es auch den Einsatz von fortgeschrittenen Schaltungskonzepten und Architekturen erfordert um das volle Potential der Technologie zur Entfaltung zu bringen.

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PART

I

Introduction & THz Integrated

Circuit Technologies

The first part of the thesis comprises two introductory chapters. The first chapter provides a brief historical context to the work presented here and outlines the motivation in reference to contemporary as well as numerous future applications. The second chapter presents a review of the most related high-frequency design priniciples and topics used throughout this work.

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CHAPTER

1 Introduction

The introduction to this thesis aims to provide a point of reference to the reader in terms of the current state-of-the-art in millimeter-wave (mm-wave) and THz integrated circuit design. It is therefore essential to provide at least a brief historical perspective, illuminating when and how the major milestones were reached and who were the most influential people of this field. It would be then easier to apprehend the future opportunities, the associated challenges and the vast potential applications aimed towards improving the contentment and security of the society. This chapter also explains the motivation and organizational overview of this thesis.

1.1 PN Junction to the Millimeter-Wave Chip: A History of

Innovation

With the level of advancement and sophistication reached today in the field of integrated circuit design, it is easy to forget its very humble beginning, and how profoundly it changed and accelerated the course of human progress. Today, integrated circuits and systems are ubiquitous. This has been made possible prominently due to ever decreasing cost, form factor and power consumption, a feat that could not be accomplished at the time of electromechanical relays and vacuum tube amplifiers, which were slow, big, bulky, highly power inefficient and had the tendency to burn out (much similar to the incandescent lights compared to the today’s light emitting diode (LED) bulbs). It was not until the invention of the transistor appropriately called as the “nerve cell”, that ushered in a new era of the Information Age [1–3]. It is interesting to note that the first applications which led to the prominence of silicon (Si) and germanium (Ge) as the favored materials (as against to copper oxide and selenium−cadmium), were crystal rectifiers for radar receivers working in the gigahertz range (corresponding to few centimeters of wavelength) during the World War II, in order to replace the thermionic vacuum tubes which worked well up to only several hundred megahertz [4]. When in 1940, Russel Ohl, the inventor of the p-n junction, literally shone light on a small black rod of

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1. Introduction

Si, to demonstrate a very large fluctuation of voltage, his colleagues at the Bell Telephone Laboratories were astonished to see the effect. The fluctuation was more than 10 times than had been observed in any other photocell up till that time. The piece of Si behaved as it did, because it was unwittingly cut across the boundary between purified Si and commercial Si (which contained much impurities), leading to the generation of a barrier or what it was called later on, a p-n junction, between the two regions. Without this discovery, the invention of the junction transistor, solar cells, and other semiconductor devices could not have been possible and it is because of this reason that the p-n junction can be regarded as the DNA of the Information Age. This Si-based p-n junction became the mainstay of microwave detection and encouraged physicists and scientists at Du Pont Company (around 1941) to improve the purity levels of Si, eventually reaching a level of 99.99 %, which is also undoubtedly a major milestone [5,6].

In 1945, Shockley proposed a device now called the “field-effect” transistor, consisting of a thin film of Si or Ge, whose conductivity could be controlled by an applied transverse electric field. This device however failed primarily due to surface charge carriers being trapped by the “surface state”, as reported by Bardeen [7]. The first successful indication of the “transistor effect” (the term transistor was coined by electrical engineer J. Pierce) was observed by Bardeen and Brattain on 15th December 1947, while working at the Bell Labs using a point-contact

tungsten electrode on a specially prepared n-type Ge slab with an evaporated gold plate that already had an inversion layer [8]. Although, this effect produced some voltage amplification, it was not strong enough to observe power amplification. The trick, as suggested by Bardeen, was to space the two contacts within a order of 0.05 mm. This was achieved by Brattain by wrapping a piece of gold foil around one edge of a triangular polystrene wedge and slitting the foil with razar blade, thus leading to the first ever point-contact transistor . The schematic representation is shown in Fig. 1.1(a) [9]. The transistor was able to give an overall power gain of 20 dB and worked upto a frequency of 1 kHz. It is interesting to know that the most important invention of the 20th century was conceived independently not just once, but twice, and also around the same time. Two German physicists, H. Mataré and H. Welker, while working in Paris, invented the “transistron,” most remarkably similar to the point-contact transistor.

However, Shockley recognizing the mechanical fragility of the point-contact transistor, devised a distinctly different transistor based on the p-n junction by R. Ohl. He also claimed the possiblity of minority-carrier injection in the bulk of the semiconductor (as opposed to the flow of holes predominantly through the inversion layer at the surface), and which ultimately led to the invention of the junction transistor, a month later on 23rd January 1948, shown in

Fig. 1.1(b). This junction transistor was devised in a form of two p-n junctions placed back to back, with non-rectifying electrical contacts [10, 11]. Though, the junction transistor behaved very similar to a vaccuum-tube triode, it had almost none of its short comings, it being smaller, more reliable and power efficient. Thus, it was the beginning of the modern semiconductor era, 4

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1.1 PN Junction to the Millimeter-Wave Chip: A History of Innovation

n-type germanium block p-type inversion layer metal plate Base Emitter Collector VEB RL VBC ∼0.05 mm ∼ Spring Polystyrene wedge vin _i_in _i_out

wrapped with a gold foil

(a) Base Single Ge crystal VBE RL ∼ vin iin iout Ga rich p-type Sb rich richSb n-type n-type Emitter Collector VCE (b)

Figure 1.1: (a) Schematic diagram of the first point-contact transistor. It was connected in a CB

configuration, with emitter terminal connected to a positive d.c. voltage, generating holes in a surface layer (the p-type inversion layer). The holes get swept by the negative voltage at the collector, producing an amplified signal at the load. (b) The first junction transistor (npn) biased in an amplifier configuration on a single Ge crystal.

and the path to the ultimate triumph of transistor electronics over vacuum tubes was set. J. Bardeen, W. H. Brattain and W. B. Shockley were awarded the Noble Prize in Physics 1956. Similar to the first junction transistor, all the earliest transistors were manufactured from Ge, until G. Teal introduced the first Si-based n-p-n transistor at Texas Instruments in 1954 using grown-junction technique. Though Ge was easier to work with and had higher frequency of operation, Si offered superior electrical advantages in terms of power-handling, break down voltages, higher temperature handling and especially very low leakage currents in the “off” state. The native oxide of Si (silicon dioxide: SiO2) is an excellent insulator which revolutionized

the integrated circuits. Furthermore, the mechanical properties of Si, ranging from extraction, purification and doping to its crystalline structure, all offer distinct advantages over other semiconductor materials [12, 13]. Si coincidently happens to be one of the most abundant 5

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1. Introduction

Figure 1.2:The first IC (size: 1.6 mm ×11.1 mm), containing a single transistor, several resistors and

a capacitor on a slice of Ge, interconnected using “flying wires” of gold. Picture courtesy Texas Instruments Incorporated.

element in Earth’s crust. By late 1954, transistors had reached operating frequencies up to 170 MHz, employing the diffusion process, introduced by M. Tanenbaum [14].

Most of the efforts then concentrated on improving the practicality of the transistor. Increasingly complex electronic circuits were designed, consisting of hundreds or thousands of discrete components, interconnected using hand-soldering, which ultimately led to a problem, called at that time as “the tyranny of numbers” [15]. The first idea to solve this problem was envisaged by G. W. A. Dummer in 1952, that by utilizing layers of amplifying, insulating, conducting and rectifying materials, all electronics could be made as a single block, avoiding the use of connecting wires [16]. The idea, however could not be realized successfully in terms of a working circuit.

In 1957, the concept of a heterojunction bipolar transistor (HBT) was first presented by Herbert Kroemer in his two landmark papers [17, 18]. The main objective was to achieve high-frequency performance by using an emitter with a higher band gap than the base region, resulting in suppressing the hole injection from the base into the emitter to increase the dc current gain, and minimizing the injection deficit. Kroemer suggested the use of Si and Ge to form an the emitter/base heterojunction. Kroemer’s Nobel Prize worthy idea1, however was

much ahead of his time, and the first practical realization was only made possible in the late 1980s, and currently a large industry has built up around the devices envisioned by Kroemer in the early years [19].

In mid 1958, J. Kilby working alone in a deserted lab at Texas Instruments®, came up with

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Herbert Kroemer, along with Zhores I. Alferov was awarded the Nobel Prize in Physics for developing semiconductor heterostructures used in high-speed- and opto-electronics, in 2000. The other co-recipient was Jack Kilby.

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1.1 PN Junction to the Millimeter-Wave Chip: A History of Innovation

an idea of micro-miniaturization, which later came to be known as “The Monolithic Idea.” He suggested making all discrete components such as resistors, capacitors and obviously transistors and diodes, in situ using a single piece of material. His supervisor at that time, W. Adcock though enthusiastic of the idea was skeptical that it would work. Later that year Kilby demonstrated a working integrated 1.3 MHz phase-shift oscillator. This first integrated circuit (IC) which revolutionized the electronic industry once again, and earned Kilby the Noble prize in Phyiscs 2000, is shown in Fig. 1.2. The oscillator was then closely followed by a flip-flop circuit built entirely from the scratch. This “Solid Circuit” concept was finally announced to the public in March, 1959 [13].

Meanwhile, Fairchild Semiconductor® had been founded in 1957, by a group of fresh Ph.D

graduates who used to work with Shockley, at a place in San Francisco, now referred to as the “Silicon Valley.” Within a very short time, J. Hoerni, a physicist at Fairchild® developed

the planar process, named because of its flat topography as compared to the mesa transistor (so named because of its raised plateu-like structure). The planar process was patented in 1959, which proposed the use of photolithography process for selectively etching, oxidation and heat diffusion, so as to realize the transistor as a two-dimensional projection, and to leave the oxide layer on the wafer, in order to protect the sensitive p-n junctions underneath. This idea, regarded as one of the most important innovation in the history of the semiconductor industry, is the foundation of today’s semiconductor industry [20–22].

R. Noyce expanded on this idea to include complete circuits in a single monolithic Si chip, with metal leads over the oxide and thus invented the first practical structure for an IC in 1959, which could be produced commercially [23,24]. Had he been alive, he would probably have received the Noble Prize with Kilby in 2000. By 1960, headed by J. Last, co-founder of Fairchild, the first commercial junction-isolated flip-flop with four-transistor and five resistors was manufactured [25]. The first metal oxide semiconductor (MOS) transistor, based on the principle of Lilienfeld and Shockley, was devised by M. M. Atalla and D. Kahng around 1960 [26, 27]. This remarkable device, owing to its smaller size and power consumption, is still present in over 99 % of microchips produced today. Later on in 1963, at the International Solid State Circuits Conference, F. M. Wanlass and C. T. Sah presented the first complementary MOS (CMOS) circuit configuration, capable of almost zero power consumption in standby mode, a remarkable innovation, especially for the low-power highly dense digital circuits [28,29]. The first general-purpose programmable microprocessor, the 4004, was invented by T. Hoff at Intel®, and was launched to the public in 1971 with a fitting title, “Announcing A New Era in

Integrated Electronics [30].” At that time, the 4004 contained 2,300 transistors. In contrast, by 2010, an Intel® Core™ processor using a 32 nm process, contains 560 million.

The discovery of the p-n junction, the invention of the transistor, the realization of the fully-integrated circuits, the development of the MOS transistors, the ingenuity of the CMOS configuration, and the era of the microprocessor, were all historical milestones, which revolu-tionized not only the electronics industry but also the lives of the common people everywhere. 7

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1. Introduction

Thereafter, the progress of the electronics industry was exponential, and it could not have been more accurately predicted than G. Moore (co-founder of Fairchild Semiconductors and later on Intel Corportation), stating that the performance and the number of components on an IC doubles every 18−24 months with the same chip price [31]. The world wide electronics industry has grown from $29 billion to an astounding $1500 billion, in a matter of 50 years [15]. The demand is still unprecedented, which can only be met with the continued development of advanced semiconductor technologies, innovative circuit implementations and exploring new frontiers especially in terms of higher frequency bands.

1.2 Motivation to Work in the mm-Wave and the THz Regime

One of the most fundamental distinction in the world of integrated circuits lies between digital and analog (also known as linear) circuits. The work presented in this thesis focuses on the later. The first widely-used analog integrated circuit was introduced by R. Widlar and D. Talbert, in 1964. It was the operational amplifier µA702 consisting of 12 BJT and 5 resistors [32,33]. This kicked started an industry which enabled the interface to the real world, which does not run on the zeroes and ones a computer understands, but nearly infinite resolution analog signals such as sound, temperature, and electromagnetic-waves. The maximum fundamental operation frequency of RF integrated circuits has been progressively increasing since then.

1.2.1 Mm-Wave Integrated Systems: Moving into Mainstream

High-frequency integrated circuits became increasingly important with a boom in the wireless industry, with the number of mobile phone users currently growing at approximately one billion per year [34]. Around 95% of the global population is now covered with a mobile cellular network, with mobile-broadband networks (3G or above) and long-term evolution (LTE) now being available to 84% and 53% of them, as reported by the International Telecommunication Union (ITU) in 2016 [35]. It has been forecasted that the unprecedented demand for high speed wireless communications, driven mostly by smartphones, tablets, and video streaming, can not be met by incremental approaches to the current LTE or 4G networks [36]. Today’s cellular providers are facing a global bandwidth shortage in the frequency spectrum ranging between 700 MHz and 2.6 GHz, where the total spectrum bandwidth does not exceed 780 MHz. It is for this reason that 5G is intended to bring a major paradigm shift to mobile communications by moving up to the mm-wave frequencies of 28 GHz and 38 GHz [37]. An ISM band is also available at 61 GHz for 5G and other potential outdoor applications [38]. Furthermore, frequency band from 57−66 GHz is being used for multi Gigabit per second (Gbps) indoor wireless communications. For long haul point-to-point wireless Gbps links, ITU has allocated 71−76 GHz and 81−86 GHz bands known as the E-Band. Within this band a contiguous bandwidth of 10 GHz is available for utilization in the USA [39]. Fortunately, with today’s 8

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1.2 Motivation to Work in the mm-Wave and the THz Regime

commercially available semiconductor technologies, fully-integrated mm-wave systems are now quite feasible.

Another major application area of mm-wave systems is in automotive radars. Once considered to be a luxury and installed on very high-end cars2, nowadays virtually every new car is

being equipped with some type of active or passive advance driver assistance systems (ADAS). Examples of ADAS include but are not limited to ACC, blind spot detection (BSD), automatic emergency braking (AEB), emergency steer assist (ESA), cross traffic alert (CTA), lane departure warning system (LDWS), forward collision warning system (FCWS), intelligent park assist (IPA), traffic jam assist (TJA), and pedestrian detection. The automotive radar market is projected to be worth several billion US dollars within the next 5 years. Historically, the center operating frequency of automotive radar systems has been constantly changing. Starting in the mid 70’s from 10 GHz3, to 35 GHz4, to 94 GHz5 to the current predominantly used

frequency bands of 24/26 GHz and 77/79 GHz [41]. Even though moving to higher frequency bands is an engineering challenge, the advantages offered are manifolds. For the same antenna aperture size, an angular separability of more than three times can be achieved at 77 GHz as compared to 24 GHz. Furthermore, a relative bandwidth of only 5% at 77 GHz is required to utilize the full 4 GHz band, making the circuit and antenna design much easier [42]. Following the trend, automotive as well as industrial radar sensors are now being developed at 122 GHz, striving for further miniaturization, which is increasingly important as only long range radars are not sufficient enough to meet the current and the future autonomous driving requirements. More sensors are now required in the short to midrange for sensing the dynamic environment around a vehicle. With higher number of radar sensors compact packaging also becomes a much more vital issue [43].

1.2.2 The Next Leap Forward: The THz Frontier

The THz regime offers many advantages and diverse applications. The potentially immense and unregulated bandwidth available from 0.27−3 THz presents a major motivation for several-Gbps to 100-several-Gbps wireless communication systems. Wireless channels have minimum latency and are more suitable for real time systems. THz wireless systems have been envisaged for terrestrial high-capacity Tbps links over distances of greater than 1 km as well as for indoor WLAN with Gbps speed [44]. Practical communications over a distance of 20 m at a data rate of 100 Gbps have been demonstrated using a combination of THz photonics and 35 nm metamorphic high electron mobility transistor (mHEMT) technology with a power-gain cut-off frequency (fmax) of 900 GHz [45]. Researchers have recently demonstrated digital data

transmission over the 300-GHz band at a rate exceeding 100 Gbps over a single channel using

2_{The first radar based adaptive cruise control (ACC) system was introduced in 1999 in a Mercedes S-class [}₄₀_] 3

from VDO GmbH, Germany

4

from Bosch GmbH and Daimler-Benz, Germany, 1975

5_{from Phillips, 1988}

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1. Introduction

a 40 nm CMOS process [46]. In terms of space applications, theoretical models have shown that even with severe atmospheric attenuation, geostationary THz satellite links can support data rates up to 1 Tbps [47].

In addition to vast potential in the field of wireless communications, the lower-THz regime finds inherent advantages in many other diverse applications. The non-ionizing nature of THz waves and the fact that THz radiation transmits through almost anything that is not metal or liquid, makes it further useful for standoff personal security/screening, and medical diagnosis [48,49]. THz waves because of their unique “fingerprint” for each substance and specific frequency-dependent absorption and dispersion properties can be used to remotely sense the composition of a chemical compound [50,51].

Radars working above 200 GHz, because of the shorter wavelength have the potential of achieving very fine resolutions comparable even to laser measurement systems. Unlike infrared, laser, and optical sensors, radars working in the sub-terahertz region have lower attenuation characteristics in fog, rain, dust, and smoke, making all-weather operation viable. A 2 GHz license-free ISM band is present at a center frequency of 245 GHz. Furthermore, the frequency range from 275 GHz to 450 GHz is still unallocated by the ITU. At these frequencies millimeter range resolution can be achieved with micrometer precision. With currently available semicon-ductor technologies high resolution image reconstruction is possible using techniques such as synthetic aperture radar (SAR) and inverse synthetic aperture radar (ISAR). Furthermore, in imaging applications the available wide bandwidths in this frequency range has the potential to reach even sub-millimeter range-resolution [52,53].

In a nutshell, currently the semiconductor industry feels a strong urge to work at mm-wave and THz frequencies. This inclination provides possibilities to improve and enhance not only many current systems, but will also enable many diversified applications, which once could only be realized using optical systems.

1.3 Overview and Organization of this Work

The work presented in this thesis, due to its diversified contents, has been divided into five parts each with various numbers of chapters. The primary aim is to present to the reader a clear structure in which each part focusses on a somewhat-similar group of circuit blocks which can be studied quite independently. Part I, comprising of this introductory chapter (Chapter 1) and Chapter 2 which is related to THz technologies, covers more general topics and theory of high frequency integrated circuit design which although is absolute essential for understanding may be skipped by the more adept readers. Part II covers three chapters in total, two of which (Chapter 3 & Chapter 4) are focussed on broadband amplifier designs each with a minimum 3-dB bandwidth of 80 GHz, while the third chapter in this part (Chapter 5) presents analysis, design and measurement results of a modified Marchand balun with improved balance and 10

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1.3 Overview and Organization of this Work

lower insertion loss over a very wide bandwidth. This type of balun has been utilized throughout this work. Part III comprising two chapters, concentrates on fundamental (Chapter 6) and subharmonic (Chapter 7) frequency down-conversion circuits. Mm-wave and THz signal sources are the subject of Part IV which comprises of three more chapters. Chapter 8 and Chapter 9 present fundamental and push-push VCOs, respectively while Chapter 10 focusses on a THz source based on a push-push frequency multiplier. The last part of this thesis, Part V primarily consisting of Chapter 11, aims to bring together many of the circuits presented in earlier chapters in a form of a 240-GHz FMCW radar sensor. In addition, it also focusses on a newly designed wideband bow-tie antenna in eWLB package. Finally, this work culminates with Chapter 12 which presents concluding remarks and proposes prospective topics and projects for the future.

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CHAPTER

2 Advanced Integrated Circuit Technologies

for Millimeter-Wave & THz Applications

2.1 A Short Comparison of III-V and Si/SiGe-Based

Semiconductor Technologies and Devices

In Chapter 1, it was argued that there is a strong need of mm-wave and THz circuits and systems to meet the demands of current and future applications. The question which naturally follows is quite straightforward: Which type of device (unipolar or bipolar) and which semiconductor integrated circuit technology (III-V or Si/silicon germanium (SiGe)) holds the most potential? The answer however is both tricky and contentious, and requires an unbiased and circumspect approach. Fig. 2.1 shows the major device families currently being used and their respective material systems1. The selection depends strongly on the potential circuit application, each

warranting a different performance metric. Economic reasons also play a vital role in shaping and reshaping the technology roadmap. Historically, III-V could not be rivaled by Si, in terms of usual needs of the semiconductor industry such as lower noise and higher output power, gain and efficiency. However, Si technology have matured significantly and up till recently have maintained their historical scaling path, and are now able to stand up to their competition [54]. III-V semiconductors because of their larger bandgaps provide higher breakdown voltages and hence are best for designing power devices2. Moreover the difference in electron mobility

between III-V and Si is very stark3.

For instance, gallium nitride (GaN) at RF and microwave frequencies provides excellent power

1

Acronyms used in Fig. 2.1 are: oxide-semiconductor field-effect transistor (MOSFET), metal-semiconductor field-effect transistor (MESFET), high-electron-mobility transistor (HEMT), pseudomorphic HEMT (p-HEMT), metamorphic HEMT (m-HEMT), silicon on insulator (SOI), laterally diffused MOSFET (LDMOS), double diffused MOSFET (DMOS), fully-depleted silicon-on-insulator (FD-SOI), Fin field effect

transistor (FinFET).

2

GaN has a bandgap of 3.39 eV as compared to Si’s 1.12 eV [55].

3_{Electron mobility of InGaAs is 11,000 cm}2_{/Vs, whereas Si has a mobility of 1300 cm}2_{/Vs [}₅₅_].

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2. Advanced Integrated Circuit Technologies for Millimeter-Wave & THz Applications

Unipolar devices Bipolar devices

MOSFET MESFET

LDMOS DMOS

FD-SOI

Si, SiGe GaAs GaAs/InGaAs

InP/InGaAs AlGaN/GaN SiGe Si GaAs InP Si/SiGe AlGaN/GaN GaAs/InGaAs Si

Solid-state electronic devices

HEMT BJT HBT

SOI

FinFET nanowire FET p-HEMT m-HEMT

Figure 2.1:Solid-state electronic device families. Most popular material systems in which the different

devices are implemented are indicated in blue. Updated from [56].

Table 2.1: Semiconductor technologies used in an iPhone7

Component/Function Technology

system-on-chip (SoC) 16 nm FinFET

Multi-band, multi-mode power amplifier (PA) InGaP/GaAs

PA control block CMOS

RF switches GaAs pHEMT

Power management ICs 28 nm CMOS

Flash memory 15 nm NAND

added efficiency (PAE) at unprecedented power levels possible in solid-state devices [57]. For achieving higher frequencies GaN on Si process has also been demonstrated. However, GaN is a difficult material to work with and integration ability is even worse than GaAs. In order to get an idea of which semiconductor technologies are being used in today’s modern devices, consider the teardown of iPhone 7 components in Table 2.1. This table shows that how scaled CMOS is dominating this segment of the market, and that state-of-the-art devices such as the FD-SOI and FinFET are already in mainstream commercial applications. For reference, a comparison of different state-of-the-art technologies for a radio-frequency integrated circuit (RFIC) is presented in Table 2.2 [58]. It is evident from the table, that as for now SiGe has surpassed both Si-based BJT and CMOS in almost all of the performance metrics, except the cost. However, as highly scaled CMOS has shown its complete dominance over the digital world, it has a strong potential to overcome its short comings in the high-frequency and mm-wave regime and challenge not only SiGe but perhaps III-V devices as well [58].

In comparison to photonics-based alternatives working in the THz range, III-V semiconductors 14

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2.2 High-Frequency Figures of Merit

Table 2.2: Performance Comparison of III-V and Si/SiGe-Based Semiconductor Technologies for

RFICs [58]

Performance SiGe Si Si III-V III-V III-V

metric HBT BJT CMOS MESFET HBT HEMT

Frequency response Very good Good Good Very good Excellent Excellent

1/f and phase noise Excellent Good Fair Poor Good Poor

Broadband noise Very good Good Good Very good Very good Excellent Linearity Very good Very good Very good Excellent Very good Excellent Power dissipation Excellent Very good Fair Fair Very good Good CMOS integration Very good Very good N/A Poor Poor Poor

IC cost Good Good Very good Fair Fair Poor

such as indium phosphide (InP) [59], indium arsenide (InAs), [60], indium gallium arsenide (InGaAs) [61], and advanced Si/SiGe [62] based technologies have shown an enormous potential in terms of feasibility, performance, and lower cost [63]. Recently, 30-nm InP HEMT devices reaching fT and fmax4 beyond 0.6 THz and 1.2 THz, respectively, have been reported [64],

with InP DHBTs not lagging much behind at fT and fmax beyond 0.52 THz and 1.1 THz,

respectively [65]. Amplification at 1 THz using 25-nm InP HEMT process has been demonstrated recently [59]. SiGe technology on the other hand has already shown its significant potential in conquering the best of both the III-V and Si worlds: achieving high performance at large scale volume and integration levels, while keeping cost to a minimum. How this feat has been achieved for SiGe-based technologies will be explained in the forthcoming sections.

2.2 High-Frequency Figures of Merit

Before delving into physics of high-frequency SiGe HBT transistor, it is important to discuss the figures of merit (FoMs) for characterizing high-frequency devices. The most commonly used FoMs are:

• unity current gain or cut-off frequency (fT),

• maximum oscillation frequency (fmax),

• minimum noise figure (NFmin),

• ring oscillator gate delay (τd), and

• transconductance efficiency (gm/IC).

For high-speed digital applications, the following FoMs are of more interest:

4_{to be defined in Section 2.2}

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2. Advanced Integrated Circuit Technologies for Millimeter-Wave & THz Applications log β(ω) fmax 1 log f JC fT JC,optfT fmax JC,optfmax NFmin JC,optNFmin MAG(f ) fT log β(ω) fmax 1 log f JC fT JC,optfT fmax JC,optfmax NFmin JC,optNFmin MAG(f ) fT

Figure 2.2:Variation of small-signal common-emitter (CE) current gain (β(ω)) and maximum available

gain (MAG) as a function of frequency and the definitions of cut-off frequency (fT) and

maximum oscillation frequency fmax. In practical cases, MAG(f) does not exhibit a region

of constant slope and cannot be accurately extrapolated. Instead, unilateral gain (U(f)) is used, which typically has a very close x-axis intercept to MAG(f).

• intrinsic slew rate (SLi), and

• maximum operation frequency of static dividers

Besides the aforementioned FoMs, other RF FoMs have also been introduced, however with much less success. For instance in [66], the authors introduce three FoMs for digital and analog RF circuits, which can be extracted directly from Y -parameter measurements and can be measured directly without involving any extrapolation. These FoMs are:

• available bandwidth of an emitter-coupled pair (fA),

• bandwidth of a cascode for a dc gain of 10 (fcasc), and

• frequency at which input admittance of a cross-coupled pair crosses zero (fcross). 2.2.1 The Unity Current Gain or Cut-off Frequency

One of the most important high-frequency parameter of a bipolar transistor is the frequency at which the common emitter (CE) small-signal current gain (β(ω)) of the transistor drops to unity, also known as the cut-off frequency or the transit frequency. The cut-off frequency can be obtained by measured or simulated Y -parameters of the transistor as follows

H21(f) =

Y21(f)

Y11(f)

(2.1) |H₂₁(f = f_T)| = 1 or 20 log|H₂₁(f = f_T)| = 0 (2.2) As evident from (2.1), fT does not depend upon the output impedance of the transistor. The

fT of advanced HBTs is often much higher than what could be measured by the available

equipments and therefore fT is usually extracted by linear extrapolation as the x-intercept

of 20 log|H21(f)| measured curves. This is illustrated in Fig. 2.2. Knowledge of fT is vital

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2.2 High-Frequency Figures of Merit gmvπ rπ Cπ Cµ ib ic B C E

Figure 2.3:High-frequency equivalent small-signal hybrid π-model for calculating f_T. before selection of a suitable technology because beyond this frequency the device is no longer useful for amplifying or for switching. In fact, most fundamental circuits are designed to be working at least below fT/3, which already requires special and careful circuit techniques and

implementation.

An expression for fT can be calculated by considering the small-signal hybrid π−model, as

shown in Fig. 2.3, with a short-circuit applied to the output [67]

ic= gmvπ − jωCµvπ, (2.3) ib = vπ 1 rπ + jωCπ+ jωCµ . (2.4)

The small-signal frequency dependent CE current gain, thus can be written as

β(ω) = ic ib = gm− jωCµ 1 rπ + jωCπ+ jωCµ . (2.5)

For frequencies of practical interest, gm  jωCµ and the equation can be simplified to

β(ω) = β0

1 + jωrπ(Cπ+ Cµ)

, (2.6)

where β0 is the small-signal low-frequency current gain and is given by

β0= gmrπ. (2.7)

At high frequencies the second term in (2.6) is much larger with respect to unity and β(ω) can be simplified to

|β(ω)| = β0

ωrπ(Cπ+ Cµ)

. (2.8)

The frequency at which the current gain |β(ω)| becomes unity, gives the following expression for ωT,

ωT =

gm

(Cπ+ Cµ)

. (2.9)

Using 2.9 and 2.6 gives a very useful and commonly used expression for the frequency-dependent 17

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2. Advanced Integrated Circuit Technologies for Millimeter-Wave & THz Applications gmvπ rπ Cπ Cµ ib ro B C E ic rB rC rE CCE

Figure 2.4:High-frequency equivalent small-signal hybrid π-model including the series parasitic

resistances, Early resistance, and the collector-emitter capacitance. This model, although still very simple, is comparatively accurate for calculating power gain and input impedance at mm-wave frequencies. small-signal gain β(ω) = β0 1 + j ω ωTβ0 . (2.10)

An approximate expression of gain-bandwidth product (GBW), valid in the roll-off region can be calculated using (2.9) and (2.10) and is given by

|β(ω)ω| ≈ ω_T. (2.11)

Equations (2.9) and (2.10) are for intrinsic fT and do not consider the effects of series resistive

parasitics which at mm-wave frequencies play a significant role. The input impedance and the power gain of the transistors in particular become strongly dependent on the emitter and base resistance as the frequency increases. The series parasitics including the transistor’s output resistance and collector-emitter capacitance are shown in Fig. 2.4.

An expression of fT which considers the effect of parasitic resistances can be calculated

following a similar approach and is given by [68] 1

ωT

= Cπ+ Cµ

gm

+ (rE+ rC)Cµ. (2.12)

The second term in (2.12) accounts for around 15−20 % of the intrinsic fT in advanced HBTs.

The contribution of other higher-order terms ignored in this calculation is usually less than 5 %. An important point to be noted here is that fT is independent of the base resistance rB. 2.2.2 Unity Power Gain Frequency or Maximum Oscillation Frequency

Before defining the unity power gain or the maximum oscillation frequency (fmax) for

high-frequency circuits, it is important to briefly discuss the different power gain definitions 18

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2.2 High-Frequency Figures of Merit gmvπ rπ Cπ Cµ rB Zout vtest itest ZS + −

Figure 2.5: Calculation of high-frequency output impedance based on an equivalent small-signal

hybrid π-model for derivation of fmax.

which depend upon the source impedance ZS and/or load impedance ZL. Three definitions

are commonly used, 1) Power gain or operating power gain G5; 2) available gain G

A6; and

3) transducer gain GT7. When both the input and the output are conjugately matched, G,

GA and GT become equal to each other and maximized, and are collectively known as the

maximum available power gain (MAG). The MAG can be calculated using Y -parameters as follows

MAG = |Y21|

|Y12|(K −

p

K2₋1), (2.13)

where K is known as the Rollet’s stability factor. The frequency at which the MAG of the transistor becomes equal to 1 is defined as fmax

MAG(f = fmax) = 1. (2.14)

As illustrated in Fig. 2.2, fmaxcan be extracted as the x-intercept of MAG plotted as a function

of frequency. In order to find an expression for fmax, it is required to find the power gain of

the transistor under the condition when both source and load are conjugately matched for which both input and output impedances must be calculated. A test setup to determine the output impedance is shown in Fig. 2.5, so that

Zout = vtest ttest , (2.15) where, itest= gmvπ + iµ. (2.16) Assuming iµ gmvπ, vπ ≈ Cµ Cπ+ Cµ vtest, (2.17) 5

G is defined by the ratio of power delivered to the load to the power delivered to the amplifier input, G = PL Pin

.

6

GA is defined by the ratio of power available from the output to the power available from the source, GA=Pavl,out

Pavl,s

.

7_GT _{is defined by the ratio of power delivered to the load to the power available from the source, G} T=

PL Pavl,s

.

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2. Advanced Integrated Circuit Technologies for Millimeter-Wave & THz Applications gmvπ rB ZS ZL Zout vS iS iin io Zin ≈ rB Zout= ZL io iout

Figure 2.6:Calculation of high-frequency output impedance based on an equivalent small-signal

hybrid π-model for derivation of fmax.

which leads to

Zout =

Cπ+ Cµ

gmCµ

(2.18) Now, the input impedance Zin is given by

Zin= rB+

1

jωCπ

(2.19) At frequencies close to fmax, the input power is mainly due to the power consumption at the

base resistance and the second term in (2.19) can be ignored. The power gain under matched condition can be calculated from

MAG(f) = 12i 2 oZL 1 2i2SZin . (2.20)

Considering Fig. 2.6, which shows that under matched condition Zin≈ rB and Zout = ZL

MAG(f) = 1 4 _i out iin 2 _Z L rB . (2.21)

Using the definition of fT given in (2.10), gives

MAG(f) = 1₄ fT f !2 ZL rB . (2.22)

At fmax, MAG becomes equal to 1, leading to

fmax= 1 2fT s ZL rB . (2.23)

Finally, substituting (2.18) leads to an approximate expression of fmax

fmax≈ s fT 8πCµrB . (2.24) 20

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2.2 High-Frequency Figures of Merit gme−jωτn rπ Cπ Cµ ro B C E rB rC rE hv2 nbi hv2 nei hi2 nci hi2 nbi

Figure 2.7: Simplified noise equivalent circuit of a high-frequency bipolar transistor. Noise sources are

shown in gray.

This equation shows that fmaxclosely follows the profile of fT and unlike fT it is also dependent

on the base resistance.

2.2.3 Minimum Noise Figure

Accurate and reliable high-frequency noise modelling of HBTs is essential for designing low-noise amplifiers (LNAs) and frequency down converters. One of the reason for SiGe HBT technology being so popular is because of its low noise capability [58]. The main sources of RF noise in a SiGe HBT are the noises associated with the dc base and collector currents and the thermal noise generated at the base resistance [69]. A simplified high-frequency noise equivalent circuit of HBT is presented in Fig. 2.7. Thermal noise sources are present at the base and the emitter, hv2

nbi and hvne2 i, respectively. The input noise current source is dominated by the base current

shot noise, hi2

nbi, and the output noise current source by the collector current shot noise, hi2nci.

At low frequencies the correlation between the base and collector noise sources is usually ignored, without loss of accuracy. However at higher frequency this assumption leads to an overestimation of transistor noise figure [70]. At mm-waves a more convenient yet accurate method for extracting circuit level noise parameters from measured two-port parameters is to employ y-parameter based noise models which take into account this correlation, such as presented in [71]. To have an understanding of the dependencies of these paramaters and how they can be effectively influenced, approximate expression for the noise parameters are provided here for reference [72]

Rn≈ rB+ rE+

n2_FVT

2IC

, (2.25)

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2. Advanced Integrated Circuit Technologies for Millimeter-Wave & THz Applications

YS,opt= GS,opt+ BS,opt≈

f fTRn    v u u t IC 2VT (rE + rB) 1 + f2 T β(ω)f2 ! + n2FfT2 4β(ω)f2 − j nF 2   , (2.26) and NFmin ≈1 + nF β(ω) + f fT v u u t2IC VT (rE + rB) 1 + f2 T β(ω)f2 ! + n2FfT2 β(ω)f2, (2.27)

where, Rn is the noise resistance, YS,opt is the optimum or noise matching source admittance,

and nF is the collector current ideality factor nearly equal to 1. As it is seen from (2.25), Rnis

independent of frequency and dependent upon the parasitic base and emitter resistances. This has been confirmed via many experiments such as demonstrated in [73]. Examining (2.26), it is noticed that the imaginary part of YS,opt is negative, requiring a series inductance at the

base for noise matching of the imaginary part. GS,opt is proportional to the frequency and the

dc collector current. Similarly, it is observed that the Fmin increases linearly with frequency

with a slope that is inversely proportional to fT [68]. Consequently, the degradation of Fmin

with frequency is slower for transistors exhibiting a higher fT. It is for this reason, it is always

desirable to keep the operating frequency of a system at least less than fT/3. Finally, for a

source termination admittance of YS = GS+ jBS, the noise figure (NF) is given by [74]

NF = NFmin+

Rn

GS

|YS− YS,opt|2. (2.28)

It is clear that when the source is noise matched YS = YS,opt, NF reaches its minimum value,

while Rn determines its sensitivity to YS,opt. Thus it is desirable to have a technology with

smallest possible NFminand Rnfor components and systems demanding low-noise performance.

2.3 Technology Aspects of the High-Frequency SiGe HBT

Transistor

As mentioned earlier, the focus of this thesis is the design of circuits and systems working in the mm-wave and THz frequency range. Most of the circuits in this work are designed to operate at frequencies even above than fT/2. At these frequencies it becomes vital to consider the

technological aspects, modelling approximations, and the physical limitations of the transistor during the circuit design phase. These topics are typically ignored for low frequency designs.

2.3.1 Intrinsic Transit Delays in a SiGe HBT

An extremely important paramater which defines a physical limit to the switching speed and maximum frequency of operation of a bipolar junction transistor (BJT) is the forward transit time τF. It is the optimization of this parameter that helps SiGe to overcome the low

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2.3 Technology Aspects of the High-Frequency SiGe HBT Transistor

HBT BJT

Emitter Base Collector

n+Si p-Si1−xGex n+Si ∆Eg,grade linear grade EC EF EV e− h+ Ge content

Figure 2.8: A linearly graded Ge base profile for a SiGe HBT. Also shown are the energy band

diagrams of a Si BJT and a SiGe HBT. The Ge grading results in a larage built-in pseudopotential which greatly reduces the base transit time for electrons.

carrier mobility and limited saturation velocity of both electrons and holes in Si. τF determines

how fast the carriers can be transported through the device under sustainable operating voltages [58]. It models the excess charge stored in a BJT under forward active operation, and is the summation of individual delay times in different regions of the transistor [67]

τF = τE+ τEBD+ τB+ τCBD, (2.29)

where τE, τEBD, τB and τCBD are the delay times associated with excess minority carrier charge

in the neutral emitter, emitter-base depletion region, and the base and the collector-base depletion region, respectively. Furthermore, the total delay also consists of the parasitic RC delays. An insight into these delay times helps to understand and minimize the overall transit time and hence improve the intrinsic high-frequency performance of the transistor.

Base Transit Time The base transit time τB is the most important component of τF

and typically determines the overall transit time. For a constant doping profile, electrons must diffuse across the base region before being swept by the collector-base electric field. By introducing a graded-base Ge profile, a drift field is induced which results in quickly accelerating the electrons to their saturation velocity, vsat. This significantly improves the

frequency response of the BJT. A typical linear-graded Ge base profile is shown in Fig.2.8. The effect of this grading on the energy band diagram is also shown in comparison with the energy band diagram of a Si BJT . The introduction of Ge in the base reduces the bandgap with respect to the bandgap of Si used in the emitter. Thus, the base transit time is strongly dependent on the minority carrier profile in the base. By definition, for a SiGe HBT with a 23

SiGe-Based Broadband Integrated Circuits and Systems for Millimeter- Wave & THz Radar Applications / submitted by Faisal Ahmed

SiGe-Based Broadband Integrated

Circuits & Systems for

Millimeter-Wave & THz Radar Applications

Doctoral Thesis

Doktor der technischen Wissenschaften

Dissertation

SiGe-Based Broadband

Integrated Circuits and Systems

for Millimeter-Wave & THz

Radar Applications

Faisal Ahmed

Institute for Communications Engineering and RF-Systems

Johannes Kepler University, Linz, Austria

Statutory Declaration

Acknowledgments

Abstract

Kurzfassung

Table of Contents

PART

I

Introduction & THz Integrated

Circuit Technologies

1

Introduction

1.1 PN Junction to the Millimeter-Wave Chip: A History of

Innovation

1.2 Motivation to Work in the mm-Wave and the THz Regime

1.3 Overview and Organization of this Work

2

Advanced Integrated Circuit Technologies

for Millimeter-Wave & THz Applications

2.1 A Short Comparison of III-V and Si/SiGe-Based

Semiconductor Technologies and Devices

2.2 High-Frequency Figures of Merit

2.3 Technology Aspects of the High-Frequency SiGe HBT

Transistor